Object identification system and method

ABSTRACT

An object identification system includes a guided surface waveguide probe that produces a guided surface wave; and an object identification tag having a receive structure and a tag circuit, the tag circuit coupled to the receive structure and electrically powered as a load on the probe by conversion of the guided surface wave to electrical current at the receive structure, the tag circuit configured to emit a return signal containing a tag identifier when electrically powered by presence of the guided surface wave.

RELATED APPLICATION DATA

This application is related to co-pending U.S. Non-provisional patent application entitled “Excitation and Use of Guided Surface Wave Modes on Lossy Media,” which was filed on Mar. 7, 2013 and assigned application Ser. No. 13/789,538, and was published on Sep. 11, 2014 as Publication Number US2014/0252886 A1, and which is incorporated herein by reference in its entirety. This application is also related to co-pending U.S. Non-provisional patent application entitled “Excitation and Use of Guided Surface Wave Modes on Lossy Media,” which was filed on Mar. 7, 2013 and assigned application Ser. No. 13/789,525, and was published on Sep. 11, 2014 as Publication Number US2014/0252865 A1, and which is incorporated herein by reference in its entirety. This application is further related to co-pending U.S. Non-provisional patent application entitled “Excitation and Use of Guided Surface Wave Modes on Lossy Media,” which was filed on Sep. 10, 2014 and assigned application Ser. No. 14/483,089, and which is incorporated herein by reference in its entirety. This application is further related to co-pending U.S. Non-provisional patent application entitled “Excitation and Use of Guided Surface Waves,” which was filed on Jun. 2, 2015 and assigned application Ser. No. 14/728,507, and which is incorporated herein by reference in its entirety. This application is further related to co-pending U.S. Non-provisional patent application entitled “Excitation and Use of Guided Surface Waves,” which was filed on Jun. 2, 2015 and assigned application Ser. No. 14/728,492, and which is incorporated herein by reference in its entirety.

BACKGROUND

For over a century, radio wave signals have been transmitted using conventional antenna structures. In contrast to radio science, electrical power distribution has relied on guiding electrical energy along electrical conductors such as wires. This understanding of the distinction between radio frequency (RF) and power transmission has existed since the early 1900's.

Radio frequency identification (RFID) systems, however, have used RF energy that is emitted from a reader device to power tags. The tags may affect the emitted signal to invoke a change in the emitted signal that is detectable by the reader device or the tags may transmit an RF signal that is detectable by the reader device. In the former case, the reader may be able to determine that a tag is within an operable range of the reader device. In the later case, the reader may be able to extract a code that uniquely identifies the tag from the signal output by the tag. The range of RFID systems is severely limited. Also, the capabilities of the tags are limited due to the small amount of useable energy that may be derived from the RF signal emitted by the reader device.

BRIEF DESCRIPTION OF THE DRAWINGS

Aspects of the present disclosure are better understood with reference to the following drawings. The drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the disclosure. Moreover, in the drawings, like reference numerals designate corresponding parts throughout the several views.

FIG. 1 is a chart that depicts field strength as a function of distance for a guided electromagnetic field and a radiated electromagnetic field.

FIG. 2 is a drawing that illustrates a propagation interface with two regions employed for transmission of a guided surface wave according to various embodiments of the present disclosure.

FIG. 3 is a drawing that illustrates a guided surface waveguide probe disposed with respect to a propagation interface of FIG. 2 according to various embodiments of the present disclosure.

FIG. 4 is a plot of an example of the magnitudes of close-in and far-out asymptotes of first order Hankel functions according to various embodiments of the present disclosure.

FIGS. 5A and 5B are drawings that illustrate a complex angle of incidence of an electric field synthesized by a guided surface waveguide probe according to various embodiments of the present disclosure.

FIG. 6 is a graphical representation illustrating the effect of elevation of a charge terminal on the location where the electric field of FIG. 5A intersects with the lossy conducting medium at a Brewster angle according to various embodiments of the present disclosure.

FIG. 7 is a graphical representation of an example of a guided surface waveguide probe according to various embodiments of the present disclosure.

FIGS. 8A through 8C are graphical representations illustrating examples of equivalent image plane models of the guided surface waveguide probe of FIGS. 3 and 7 according to various embodiments of the present disclosure.

FIGS. 9A and 9B are graphical representations illustrating examples of single-wire transmission line and classic transmission line models of the equivalent image plane models of FIGS. 8B and 8C according to various embodiments of the present disclosure.

FIG. 10 is a flow chart illustrating an example of adjusting a guided surface waveguide probe of FIGS. 3 and 7 to launch a guided surface wave along the surface of a lossy conducting medium according to various embodiments of the present disclosure.

FIG. 11 is a plot illustrating an example of the relationship between a wave tilt angle and the phase delay of a guided surface waveguide probe of FIGS. 3 and 7 according to various embodiments of the present disclosure.

FIG. 12 is a drawing that illustrates an example of a guided surface waveguide probe according to various embodiments of the present disclosure.

FIG. 13 is a graphical representation illustrating the incidence of a synthesized electric field at a complex Brewster angle to match the guided surface waveguide mode at the Hankel crossover distance according to various embodiments of the present disclosure.

FIG. 14 is a graphical representation of an example of a guided surface waveguide probe of FIG. 12 according to various embodiments of the present disclosure.

FIG. 15A includes plots of an example of the imaginary and real parts of a phase delay (Φ_(U)) of a charge terminal T₁ of a guided surface waveguide probe according to various embodiments of the present disclosure.

FIG. 15B is a schematic diagram of the guided surface waveguide probe of FIG. 14 according to various embodiments of the present disclosure.

FIG. 16 is a drawing that illustrates an example of a guided surface waveguide probe according to various embodiments of the present disclosure.

FIG. 17 is a graphical representation of an example of a guided surface waveguide probe of FIG. 16 according to various embodiments of the present disclosure.

FIGS. 18A through 18C depict examples of receiving structures that can be employed to receive energy transmitted in the form of a guided surface wave launched by a guided surface waveguide probe according to the various embodiments of the present disclosure.

FIG. 18D is a flow chart illustrating an example of adjusting a receiving structure according to various embodiments of the present disclosure.

FIG. 19 depicts an example of an additional receiving structure that can be employed to receive energy transmitted in the form of a guided surface wave launched by a guided surface waveguide probe according to the various embodiments of the present disclosure.

FIG. 20A shows a symbol that generically represents a guided surface wave waveguide probe.

FIG. 20B shows a symbol that generically represents a guided surface wave receive structure.

FIG. 20C shows a symbol that generically represents a linear probe type of guided surface wave receive structure.

FIG. 20D shows a symbol that generically represents a tuned resonator type of guided surface wave receive structure.

FIG. 20E shows a symbol that generically represents a magnetic coil type of guided surface wave receive structure.

FIG. 21 is a schematic illustration of one embodiment of an object identification system.

FIG. 22 is a schematic illustration of another embodiment of an object identification system.

FIG. 23 is a schematic illustration of a tag that is used as part of the object identification system.

FIG. 24 is a schematic view of first and second object identification systems deployed at neighboring sites.

FIG. 25 is a schematic view of an object identification system deployed to identify objects over a wide area.

FIG. 26 is a schematic illustration of a computer system and a receiver that are used as part of the object identification system.

DETAILED DESCRIPTION 1. Surface-Guided Transmission Line Devices and Signal Generation

To begin, some terminology shall be established to provide clarity in the discussion of concepts to follow. First, as contemplated herein, a formal distinction is drawn between radiated electromagnetic fields and guided electromagnetic fields.

As contemplated herein, a radiated electromagnetic field comprises electromagnetic energy that is emitted from a source structure in the form of waves that are not bound to a waveguide. For example, a radiated electromagnetic field is generally a field that leaves an electric structure such as an antenna and propagates through the atmosphere or other medium and is not bound to any waveguide structure. Once radiated electromagnetic waves leave an electric structure such as an antenna, they continue to propagate in the medium of propagation (such as air) independent of their source until they dissipate regardless of whether the source continues to operate. Once electromagnetic waves are radiated, they are not recoverable unless intercepted, and, if not intercepted, the energy inherent in the radiated electromagnetic waves is lost forever. Electrical structures such as antennas are designed to radiate electromagnetic fields by maximizing the ratio of the radiation resistance to the structure loss resistance. Radiated energy spreads out in space and is lost regardless of whether a receiver is present. The energy density of the radiated fields is a function of distance due to geometric spreading. Accordingly, the term “radiate” in all its forms as used herein refers to this form of electromagnetic propagation.

A guided electromagnetic field is a propagating electromagnetic wave whose energy is concentrated within or near boundaries between media having different electromagnetic properties. In this sense, a guided electromagnetic field is one that is bound to a waveguide and may be characterized as being conveyed by the current flowing in the waveguide. If there is no load to receive and/or dissipate the energy conveyed in a guided electromagnetic wave, then no energy is lost except for that dissipated in the conductivity of the guiding medium. Stated another way, if there is no load for a guided electromagnetic wave, then no energy is consumed. Thus, a generator or other source generating a guided electromagnetic field does not deliver real power unless a resistive load is present. To this end, such a generator or other source essentially runs idle until a load is presented. This is akin to running a generator to generate a 60 Hertz electromagnetic wave that is transmitted over power lines where there is no electrical load. It should be noted that a guided electromagnetic field or wave is the equivalent to what is termed a “transmission line mode.” This contrasts with radiated electromagnetic waves in which real power is supplied at all times in order to generate radiated waves. Unlike radiated electromagnetic waves, guided electromagnetic energy does not continue to propagate along a finite length waveguide after the energy source is turned off. Accordingly, the term “guide” in all its forms as used herein refers to this transmission mode of electromagnetic propagation.

Referring now to FIG. 1, shown is a graph 100 of field strength in decibels (dB) above an arbitrary reference in volts per meter as a function of distance in kilometers on a log-dB plot to further illustrate the distinction between radiated and guided electromagnetic fields. The graph 100 of FIG. 1 depicts a guided field strength curve 103 that shows the field strength of a guided electromagnetic field as a function of distance. This guided field strength curve 103 is essentially the same as a transmission line mode. Also, the graph 100 of FIG. 1 depicts a radiated field strength curve 106 that shows the field strength of a radiated electromagnetic field as a function of distance.

Of interest are the shapes of the curves 103 and 106 for guided wave and for radiation propagation, respectively. The radiated field strength curve 106 falls off geometrically (1/d, where d is distance), which is depicted as a straight line on the log-log scale. The guided field strength curve 103, on the other hand, has a characteristic exponential decay of e^(−αd)/√{square root over (d)} and exhibits a distinctive knee 109 on the log-log scale. The guided field strength curve 103 and the radiated field strength curve 106 intersect at point 112, which occurs at a crossing distance. At distances less than the crossing distance at intersection point 112, the field strength of a guided electromagnetic field is significantly greater at most locations than the field strength of a radiated electromagnetic field. At distances greater than the crossing distance, the opposite is true. Thus, the guided and radiated field strength curves 103 and 106 further illustrate the fundamental propagation difference between guided and radiated electromagnetic fields. For an informal discussion of the difference between guided and radiated electromagnetic fields, reference is made to Milligan, T., Modern Antenna Design, McGraw-Hill, 1^(st) Edition, 1985, pp. 8-9, which is incorporated herein by reference in its entirety.

The distinction between radiated and guided electromagnetic waves, made above, is readily expressed formally and placed on a rigorous basis. That two such diverse solutions could emerge from one and the same linear partial differential equation, the wave equation, analytically follows from the boundary conditions imposed on the problem. The Green function for the wave equation, itself, contains the distinction between the nature of radiation and guided waves.

In empty space, the wave equation is a differential operator whose eigenfunctions possess a continuous spectrum of eigenvalues on the complex wave-number plane. This transverse electro-magnetic (TEM) field is called the radiation field, and those propagating fields are called “Hertzian waves.” However, in the presence of a conducting boundary, the wave equation plus boundary conditions mathematically lead to a spectral representation of wave-numbers composed of a continuous spectrum plus a sum of discrete spectra. To this end, reference is made to Sommerfeld, A., “Uber die Ausbreitung der Wellen in der Drahtlosen Telegraphie,” Annalen der Physik, Vol. 28, 1909, pp. 665-736. Also see Sommerfeld, A., “Problems of Radio,” published as Chapter 6 in Partial Differential Equations in Physics—Lectures on Theoretical Physics: Volume VI, Academic Press, 1949, pp. 236-289, 295-296; Collin, R. E., “Hertzian Dipole Radiating Over a Lossy Earth or Sea: Some Early and Late 20^(th) Century Controversies,” IEEE Antennas and Propagation Magazine, Vol. 46, No. 2, April 2004, pp. 64-79; and Reich, H. J., Ordnung, P. F, Krauss, H. L., and Skalnik, J. G., Microwave Theory and Techniques, Van Nostrand, 1953, pp. 291-293, each of these references being incorporated herein by reference in its entirety.

The terms “ground wave” and “surface wave” identify two distinctly different physical propagation phenomena. A surface wave arises analytically from a distinct pole yielding a discrete component in the plane wave spectrum. See, e.g., “The Excitation of Plane Surface Waves” by Cullen, A. L., (Proceedings of the IEE (British), Vol. 101, Part IV, August 1954, pp. 225-235). In this context, a surface wave is considered to be a guided surface wave. The surface wave (in the Zenneck-Sommerfeld guided wave sense) is, physically and mathematically, not the same as the ground wave (in the Weyl-Norton-FCC sense) that is now so familiar from radio broadcasting. These two propagation mechanisms arise from the excitation of different types of eigenvalue spectra (continuum or discrete) on the complex plane. The field strength of the guided surface wave decays exponentially with distance as illustrated by curve 103 of FIG. 1 (much like propagation in a lossy waveguide) and resembles propagation in a radial transmission line, as opposed to the classical Hertzian radiation of the ground wave, which propagates spherically, possesses a continuum of eigenvalues, falls off geometrically as illustrated by curve 106 of FIG. 1, and results from branch-cut integrals. As experimentally demonstrated by C. R. Burrows in “The Surface Wave in Radio Propagation over Plane Earth” (Proceedings of the IRE, Vol. 25, No. 2, February, 1937, pp. 219-229) and “The Surface Wave in Radio Transmission” (Bell Laboratories Record, Vol. 15, June 1937, pp. 321-324), vertical antennas radiate ground waves but do not launch guided surface waves.

To summarize the above, first, the continuous part of the wave-number eigenvalue spectrum, corresponding to branch-cut integrals, produces the radiation field, and second, the discrete spectra, and corresponding residue sum arising from the poles enclosed by the contour of integration, result in non-TEM traveling surface waves that are exponentially damped in the direction transverse to the propagation. Such surface waves are guided transmission line modes. For further explanation, reference is made to Friedman, B., Principles and Techniques of Applied Mathematics, Wiley, 1956, pp. pp. 214, 283-286, 290, 298-300.

In free space, antennas excite the continuum eigenvalues of the wave equation, which is a radiation field, where the outwardly propagating RF energy with E_(z) and H_(ϕ) in-phase is lost forever. On the other hand, waveguide probes excite discrete eigenvalues, which results in transmission line propagation. See Collin, R. E., Field Theory of Guided Waves, McGraw-Hill, 1960, pp. 453, 474-477. While such theoretical analyses have held out the hypothetical possibility of launching open surface guided waves over planar or spherical surfaces of lossy, homogeneous media, for more than a century no known structures in the engineering arts have existed for accomplishing this with any practical efficiency. Unfortunately, since it emerged in the early 1900's, the theoretical analysis set forth above has essentially remained a theory and there have been no known structures for practically accomplishing the launching of open surface guided waves over planar or spherical surfaces of lossy, homogeneous media.

According to the various embodiments of the present disclosure, various guided surface waveguide probes are described that are configured to excite electric fields that couple into a guided surface waveguide mode along the surface of a lossy conducting medium. Such guided electromagnetic fields are substantially mode-matched in magnitude and phase to a guided surface wave mode on the surface of the lossy conducting medium. Such a guided surface wave mode can also be termed a Zenneck waveguide mode. By virtue of the fact that the resultant fields excited by the guided surface waveguide probes described herein are substantially mode-matched to a guided surface waveguide mode on the surface of the lossy conducting medium, a guided electromagnetic field in the form of a guided surface wave is launched along the surface of the lossy conducting medium. According to one embodiment, the lossy conducting medium comprises a terrestrial medium such as the Earth.

Referring to FIG. 2, shown is a propagation interface that provides for an examination of the boundary value solutions to Maxwell's equations derived in 1907 by Jonathan Zenneck as set forth in his paper Zenneck, J., “On the Propagation of Plane Electromagnetic Waves Along a Flat Conducting Surface and their Relation to Wireless Telegraphy,” Annalen der Physik, Serial 4, Vol. 23, Sep. 20, 1907, pp. 846-866. FIG. 2 depicts cylindrical coordinates for radially propagating waves along the interface between a lossy conducting medium specified as Region 1 and an insulator specified as Region 2. Region 1 can comprise, for example, any lossy conducting medium. In one example, such a lossy conducting medium can comprise a terrestrial medium such as the Earth or other medium. Region 2 is a second medium that shares a boundary interface with Region 1 and has different constitutive parameters relative to Region 1. Region 2 can comprise, for example, any insulator such as the atmosphere or other medium. The reflection coefficient for such a boundary interface goes to zero only for incidence at a complex Brewster angle. See Stratton, J. A., Electromagnetic Theory, McGraw-Hill, 1941, p. 516.

According to various embodiments, the present disclosure sets forth various guided surface waveguide probes that generate electromagnetic fields that are substantially mode-matched to a guided surface waveguide mode on the surface of the lossy conducting medium comprising Region 1. According to various embodiments, such electromagnetic fields substantially synthesize a wave front incident at a complex Brewster angle of the lossy conducting medium that can result in zero reflection.

To explain further, in Region 2, where an e^(jωt) field variation is assumed and where ρ≠0 and z≥0 (with z being the vertical coordinate normal to the surface of Region 1, and ρ being the radial dimension in cylindrical coordinates), Zenneck's closed-form exact solution of Maxwell's equations satisfying the boundary conditions along the interface are expressed by the following electric field and magnetic field components:

$\begin{matrix} {{H_{2\varphi} = {{Ae}^{{- u_{2}}z}{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},} & (1) \\ {{E_{2\rho} = {{A\left( \frac{u_{2}}{j\; {\omega ɛ}_{o}} \right)}e^{{- u_{2}}z}{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},{and}} & (2) \\ {E_{2\; z} = {{A\left( \frac{- \gamma}{{\omega ɛ}_{o}} \right)}e^{{- u_{2}}z}{{H_{0}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}.}}} & (3) \end{matrix}$

In Region 1, where the e^(jωt) field variation is assumed and where ρ≠0 and z≤0, Zenneck's closed-form exact solution of Maxwell's equations satisfying the boundary conditions along the interface is expressed by the following electric field and magnetic field components:

$\begin{matrix} {{H_{1\varphi} = {{Ae}^{{- u_{1}}z}{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},} & (4) \\ {{E_{1\rho} = {{A\left( \frac{- u_{1}}{\sigma_{1} + {j\; {\omega ɛ}_{1}}} \right)}e^{u_{1}z}{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},{and}} & (5) \\ {E_{1\; z} = {{A\left( \frac{{- j}\; \gamma}{\sigma_{1} + {j\; {\omega ɛ}_{1}}} \right)}e^{u_{1}z}{{H_{0}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}.}}} & (6) \end{matrix}$

In these expressions, z is the vertical coordinate normal to the surface of Region 1 and ρ is the radial coordinate, H_(n) ⁽²⁾(−jγρ) is a complex argument Hankel function of the second kind and order n, u₁ is the propagation constant in the positive vertical (z) direction in Region 1, u₂ is the propagation constant in the vertical (z) direction in Region 2, σ₁ is the conductivity of Region 1, ω is equal to 2πf, where f is a frequency of excitation, ε_(o) is the permittivity of free space, ε₁ is the permittivity of Region 1, A is a source constant imposed by the source, and γ is a surface wave radial propagation constant.

The propagation constants in the ±z directions are determined by separating the wave equation above and below the interface between Regions 1 and 2, and imposing the boundary conditions. This exercise gives, in Region 2,

$\begin{matrix} {u_{2} = \frac{- {jk}_{o}}{\sqrt{1 + \left( {ɛ_{r} - {jx}} \right)}}} & (7) \end{matrix}$

and gives, in Region 1,

u ₁ =−u ₂(ε_(r) −jx).  (8)

The radial propagation constant γ is given by

$\begin{matrix} {{\gamma = {{j\sqrt{k_{o}^{2} + u_{2}^{2}}} = {j\frac{k_{o}n}{\sqrt{1 + n^{2}}}}}},} & (9) \end{matrix}$

which is a complex expression where n is the complex index of refraction given by

n=√{square root over (ε_(r) −jx)}.  (10)

In all of the above Equations,

$\begin{matrix} {{x = \frac{\sigma_{1}}{{\omega ɛ}_{o}}},{and}} & (11) \\ {{k_{o} = {{\omega \sqrt{\mu_{o}ɛ_{o}}} = \frac{\lambda_{o}}{2\pi}}},} & (12) \end{matrix}$

where ε_(r) comprises the relative permittivity of Region 1, σ₁ is the conductivity of Region 1, ε_(o) is the permittivity of free space, and μ_(o) comprises the permeability of free space. Thus, the generated surface wave propagates parallel to the interface and exponentially decays vertical to it. This is known as evanescence.

Thus, Equations (1)-(3) can be considered to be a cylindrically-symmetric, radially-propagating waveguide mode. See Barlow, H. M., and Brown, J., Radio Surface Waves, Oxford University Press, 1962, pp. 10-12, 29-33. The present disclosure details structures that excite this “open boundary” waveguide mode. Specifically, according to various embodiments, a guided surface waveguide probe is provided with a charge terminal of appropriate size that is fed with voltage and/or current and is positioned relative to the boundary interface between Region 2 and Region 1. This may be better understood with reference to FIG. 3, which shows an example of a guided surface waveguide probe 200 a that includes a charge terminal T₁ elevated above a lossy conducting medium 203 (e.g., the Earth) along a vertical axis z that is normal to a plane presented by the lossy conducting medium 203. The lossy conducting medium 203 makes up Region 1, and a second medium 206 makes up Region 2 and shares a boundary interface with the lossy conducting medium 203.

According to one embodiment, the lossy conducting medium 203 can comprise a terrestrial medium such as the planet Earth. To this end, such a terrestrial medium comprises all structures or formations included thereon whether natural or man-made. For example, such a terrestrial medium can comprise natural elements such as rock, soil, sand, fresh water, sea water, trees, vegetation, and all other natural elements that make up our planet. In addition, such a terrestrial medium can comprise man-made elements such as concrete, asphalt, building materials, and other man-made materials. In other embodiments, the lossy conducting medium 203 can comprise some medium other than the Earth, whether naturally occurring or man-made. In other embodiments, the lossy conducting medium 203 can comprise other media such as man-made surfaces and structures such as automobiles, aircraft, man-made materials (such as plywood, plastic sheeting, or other materials) or other media.

In the case where the lossy conducting medium 203 comprises a terrestrial medium or Earth, the second medium 206 can comprise the atmosphere above the ground. As such, the atmosphere can be termed an “atmospheric medium” that comprises air and other elements that make up the atmosphere of the Earth. In addition, it is possible that the second medium 206 can comprise other media relative to the lossy conducting medium 203.

The guided surface waveguide probe 200 a includes a feed network 209 that couples an excitation source 212 to the charge terminal T₁ via, e.g., a vertical feed line conductor. According to various embodiments, a charge Q₁ is imposed on the charge terminal T₁ to synthesize an electric field based upon the voltage applied to terminal T₁ at any given instant. Depending on the angle of incidence (θ_(i)) of the electric field (E), it is possible to substantially mode-match the electric field to a guided surface waveguide mode on the surface of the lossy conducting medium 203 comprising Region 1.

By considering the Zenneck closed-form solutions of Equations (1)-(6), the Leontovich impedance boundary condition between Region 1 and Region 2 can be stated as

{circumflex over (z)}×{right arrow over (H)} ₂(ρ,φ,0)={right arrow over (J)} _(S),  (13)

where {circumflex over (z)} is a unit normal in the positive vertical (+z) direction and {right arrow over (H)}₂ is the magnetic field strength in Region 2 expressed by Equation (1) above. Equation (13) implies that the electric and magnetic fields specified in Equations (1)-(3) may result in a radial surface current density along the boundary interface, where the radial surface current density can be specified by

J _(ρ)(ρ′)=−AH ₁ ⁽²⁾(−jγρ′)  (14)

where A is a constant. Further, it should be noted that close-in to the guided surface waveguide probe 200 (for ρ<<λ), Equation (14) above has the behavior

$\begin{matrix} {{J_{close}\left( \rho^{\prime} \right)} = {\frac{- {A\left( {j\; 2} \right)}}{\pi \left( {{- j}\; {\gamma\rho}^{\prime}} \right)} = {{- H_{\varphi}} = {- {\frac{I_{o}}{2{\pi\rho}^{\prime}}.}}}}} & (15) \end{matrix}$

The negative sign means that when source current (I_(o)) flows vertically upward as illustrated in FIG. 3, the “close-in” ground current flows radially inward. By field matching on H_(ϕ) “close-in,” it can be determined that

$\begin{matrix} {A = {{- \frac{I_{o}\gamma}{4}} = {- \frac{\omega \; q_{1}\gamma}{4}}}} & (16) \end{matrix}$

where q₁=C₁V₁, in Equations (1)-(6) and (14). Therefore, the radial surface current density of Equation (14) can be restated as

$\begin{matrix} {{J_{\rho}\left( \rho^{\prime} \right)} = {\frac{I_{o}\gamma}{4}{{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}^{\prime}} \right)}.}}} & (17) \end{matrix}$

The fields expressed by Equations (1)-(6) and (17) have the nature of a transmission line mode bound to a lossy interface, not radiation fields that are associated with groundwave propagation. See Barlow, H. M. and Brown, J., Radio Surface Waves, Oxford University Press, 1962, pp. 1-5.

At this point, a review of the nature of the Hankel functions used in Equations (1)-(6) and (17) is provided for these solutions of the wave equation. One might observe that the Hankel functions of the first and second kind and order n are defined as complex combinations of the standard Bessel functions of the first and second kinds

H _(n) ⁽¹⁾(x)=J _(n)(x)+jN _(n)(x), and  (18)

H _(n) ⁽²⁾(x)=J _(n)(x)−jN _(n)(x),  (19)

These functions represent cylindrical waves propagating radially inward (H_(n) ⁽¹⁾) and outward (H_(n) ⁽²⁾), respectively. The definition is analogous to the relationship e^(±jx)=cos x±j sin x. See, for example, Harrington, R. F., Time-Harmonic Fields, McGraw-Hill, 1961, pp. 460-463.

That H_(n) ⁽²⁾(k_(ρ)ρ) is an outgoing wave can be recognized from its large argument asymptotic behavior that is obtained directly from the series definitions of J_(n)(x) and N_(n)(x). Far-out from the guided surface waveguide probe:

$\begin{matrix} {{{{H_{n}^{(2)}(x)}\underset{x\rightarrow\infty}{\rightarrow}{\sqrt{\frac{2\; j}{\pi \; x}}j^{n}e^{- {jx}}}} = {\sqrt{\frac{2}{\pi \; x}}j^{n}e^{- {j{({x - \frac{\pi}{4}})}}}}},} & \left( {20a} \right) \end{matrix}$

which, when multiplied by e^(jωt), is an outward propagating cylindrical wave of the form e^(j(ωt−kρ)) with a 1/√{square root over (ρ)} spatial variation. The first order (n=1) solution can be determined from Equation (20a) to be

$\begin{matrix} {{{H_{1}^{(2)}(x)}\underset{x\rightarrow\infty}{\rightarrow}{j\sqrt{\frac{2\; j}{\pi \; x}}e^{- {jx}}}} = {\sqrt{\frac{2}{\pi \; x}}{e^{- {j{({x - \frac{\pi}{2} - \frac{\pi}{4}})}}}.}}} & \left( {20b} \right) \end{matrix}$

Close-in to the guided surface waveguide probe (for ρ<<λ), the Hankel function of first order and the second kind behaves as

$\begin{matrix} {{H_{1}^{(2)}(x)}\underset{x\rightarrow 0}{\rightarrow}{\frac{2\; j}{\pi \; x}.}} & (21) \end{matrix}$

Note that these asymptotic expressions are complex quantities. When x is a real quantity, Equations (20b) and (21) differ in phase by √{square root over (j)}, which corresponds to an extra phase advance or “phase boost” of 45° or, equivalently, λ/8. The close-in and far-out asymptotes of the first order Hankel function of the second kind have a Hankel “crossover” or transition point where they are of equal magnitude at a distance of ρ=R_(x).

Thus, beyond the Hankel crossover point the “far out” representation predominates over the “close-in” representation of the Hankel function. The distance to the Hankel crossover point (or Hankel crossover distance) can be found by equating Equations (20b) and (21) for −jγρ, and solving for R_(x). With x=σ/ωε_(o), it can be seen that the far-out and close-in Hankel function asymptotes are frequency dependent, with the Hankel crossover point moving out as the frequency is lowered. It should also be noted that the Hankel function asymptotes may also vary as the conductivity (σ) of the lossy conducting medium changes. For example, the conductivity of the soil can vary with changes in weather conditions.

Referring to FIG. 4, shown is an example of a plot of the magnitudes of the first order Hankel functions of Equations (20b) and (21) for a Region 1 conductivity of σ=0.010 mhos/m and relative permittivity ε_(r)=15, at an operating frequency of 1850 kHz. Curve 115 is the magnitude of the far-out asymptote of Equation (20b) and curve 118 is the magnitude of the close-in asymptote of Equation (21), with the Hankel crossover point 121 occurring at a distance of R_(x)=54 feet. While the magnitudes are equal, a phase offset exists between the two asymptotes at the Hankel crossover point 121. It can also be seen that the Hankel crossover distance is much less than a wavelength of the operation frequency.

Considering the electric field components given by Equations (2) and (3) of the Zenneck closed-form solution in Region 2, it can be seen that the ratio of E_(z) and E_(ρ) asymptotically passes to

$\begin{matrix} {{\frac{E_{z}}{E_{\rho}} = {{{\left( \frac{{- j}\; \gamma}{u_{2}} \right)\frac{H_{0}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}\underset{\rho\rightarrow\infty}{\rightarrow}\sqrt{ɛ_{r} - {j\frac{\sigma}{{\omega ɛ}_{o}}}}} = {n = {\tan \; \theta_{i}}}}},} & (22) \end{matrix}$

where n is the complex index of refraction of Equation (10) and θ_(i) is the angle of incidence of the electric field. In addition, the vertical component of the mode-matched electric field of Equation (3) asymptotically passes to

$\begin{matrix} {{E_{2\; z}\underset{\rho\rightarrow\infty}{\rightarrow}{\left( \frac{q_{free}}{ɛ_{o}} \right)\sqrt{\frac{\gamma^{3}}{8\pi}}e^{{- u_{2}}z}\frac{e^{- {j{({{\gamma\rho} - {\pi/4}})}}}}{\sqrt{\rho}}}},} & (23) \end{matrix}$

which is linearly proportional to free charge on the isolated component of the elevated charge terminal's capacitance at the terminal voltage, q_(free)=C_(free)×V_(T).

For example, the height H₁ of the elevated charge terminal T₁ in FIG. 3 affects the amount of free charge on the charge terminal T₁. When the charge terminal T₁ is near the ground plane of Region 1, most of the charge Q₁ on the terminal is “bound.” As the charge terminal T₁ is elevated, the bound charge is lessened until the charge terminal T₁ reaches a height at which substantially all of the isolated charge is free.

The advantage of an increased capacitive elevation for the charge terminal T₁ is that the charge on the elevated charge terminal T₁ is further removed from the ground plane, resulting in an increased amount of free charge q_(free) to couple energy into the guided surface waveguide mode. As the charge terminal T₁ is moved away from the ground plane, the charge distribution becomes more uniformly distributed about the surface of the terminal. The amount of free charge is related to the self-capacitance of the charge terminal T₁.

For example, the capacitance of a spherical terminal can be expressed as a function of physical height above the ground plane. The capacitance of a sphere at a physical height of h above a perfect ground is given by

C _(elevated sphere)=4πε_(o) a(1+M+M ² +M ³+2M ⁴+3M ⁵+ . . . ),  (24)

where the diameter of the sphere is 2a, and where M=a/2h with h being the height of the spherical terminal. As can be seen, an increase in the terminal height h reduces the capacitance C of the charge terminal. It can be shown that for elevations of the charge terminal T₁ that are at a height of about four times the diameter (4D=8a) or greater, the charge distribution is approximately uniform about the spherical terminal, which can improve the coupling into the guided surface waveguide mode.

In the case of a sufficiently isolated terminal, the self-capacitance of a conductive sphere can be approximated by C=4πε_(o)a, where a is the radius of the sphere in meters, and the self-capacitance of a disk can be approximated by C=8ε_(o)a, where a is the radius of the disk in meters. The charge terminal T₁ can include any shape such as a sphere, a disk, a cylinder, a cone, a torus, a hood, one or more rings, or any other randomized shape or combination of shapes. An equivalent spherical diameter can be determined and used for positioning of the charge terminal T₁.

This may be further understood with reference to the example of FIG. 3, where the charge terminal T₁ is elevated at a physical height of h_(p)=H₁ above the lossy conducting medium 203. To reduce the effects of the “bound” charge, the charge terminal T₁ can be positioned at a physical height that is at least four times the spherical diameter (or equivalent spherical diameter) of the charge terminal T₁ to reduce the bounded charge effects.

Referring next to FIG. 5A, shown is a ray optics interpretation of the electric field produced by the elevated charge Q₁ on charge terminal T₁ of FIG. 3. As in optics, minimizing the reflection of the incident electric field can improve and/or maximize the energy coupled into the guided surface waveguide mode of the lossy conducting medium 203. For an electric field (ε_(∥)) that is polarized parallel to the plane of incidence (not the boundary interface), the amount of reflection of the incident electric field may be determined using the Fresnel reflection coefficient, which can be expressed as

$\begin{matrix} {{{\Gamma_{}\left( \theta_{i} \right)} = {\frac{E_{{},R}}{E_{{},i}} = \frac{\sqrt{\left( {ɛ_{r} - {jx}} \right) - {\sin^{2}\theta_{i}}} - {\left( {ɛ_{r} - {jx}} \right)\cos \; \theta_{i}}}{\sqrt{\left( {ɛ_{r} - {jx}} \right) - {\sin^{2}\theta_{i}}} + {\left( {ɛ_{r} - {jx}} \right)\cos \; \theta_{i}}}}},} & (25) \end{matrix}$

where θ_(i) is the conventional angle of incidence measured with respect to the surface normal.

In the example of FIG. 5A, the ray optic interpretation shows the incident field polarized parallel to the plane of incidence having an angle of incidence of θ_(i), which is measured with respect to the surface normal ({circumflex over (z)}). There will be no reflection of the incident electric field when Γ_(∥)(θ_(i))=0 and thus the incident electric field will be completely coupled into a guided surface waveguide mode along the surface of the lossy conducting medium 203. It can be seen that the numerator of Equation (25) goes to zero when the angle of incidence is

θ_(i)=arctan(√{square root over (ε_(r) −jx)})=θ_(i,B),  (26)

where x=σ/ωε_(o). This complex angle of incidence (θ_(i,B)) is referred to as the Brewster angle. Referring back to Equation (22), it can be seen that the same complex Brewster angle (θ_(i,B)) relationship is present in both Equations (22) and (26).

As illustrated in FIG. 5A, the electric field vector E can be depicted as an incoming non-uniform plane wave, polarized parallel to the plane of incidence. The electric field vector E can be created from independent horizontal and vertical components as

{right arrow over (E)}(θ_(i))=E _(ρ) {circumflex over (ρ)}+E _(z) {circumflex over (z)}.  (27)

Geometrically, the illustration in FIG. 5A suggests that the electric field vector E can be given by

$\begin{matrix} {{{E_{\rho}\left( {\rho,z} \right)} = {{E\left( {\rho,z} \right)}\cos \; \theta_{i}}},{and}} & \left( {28a} \right) \\ {{{E_{z}\left( {\rho,z} \right)} = {{{E\left( {\rho,z} \right)}{\cos \left( {\frac{\pi}{2} - \theta_{i}} \right)}} = {{E\left( {\rho,z} \right)}\sin \; \theta_{i}}}},} & \left( {28b} \right) \end{matrix}$

which means that the field ratio is

$\begin{matrix} {\frac{E_{\rho}}{E_{z}} = {\frac{1}{\tan \; \theta_{i}} = {\tan \; {\psi_{i}.}}}} & (29) \end{matrix}$

A generalized parameter W, called “wave tilt,” is noted herein as the ratio of the horizontal electric field component to the vertical electric field component given by

$\begin{matrix} {{W = {\frac{E_{\rho}}{E_{z}} = {{W}e^{j\; \Psi}}}},{or}} & \left( {30a} \right) \\ {{\frac{1}{W} = {\frac{E_{z}}{E_{\rho}} = {{\tan \; \theta_{i}} = {\frac{1}{W}e^{{- j}\; \Psi}}}}},} & \left( {30b} \right) \end{matrix}$

which is complex and has both magnitude and phase. For an electromagnetic wave in Region 2, the wave tilt angle (Ψ) is equal to the angle between the normal of the wave-front at the boundary interface with Region 1 and the tangent to the boundary interface. This may be easier to see in FIG. 5B, which illustrates equi-phase surfaces of an electromagnetic wave and their normals for a radial cylindrical guided surface wave. At the boundary interface (z=0) with a perfect conductor, the wave-front normal is parallel to the tangent of the boundary interface, resulting in W=0. However, in the case of a lossy dielectric, a wave tilt W exists because the wave-front normal is not parallel with the tangent of the boundary interface at z=0.

Applying Equation (30b) to a guided surface wave gives

$\begin{matrix} {{\tan \; \theta_{i,B}} = {\frac{E_{z}}{E_{\rho}} = {\frac{u_{2}}{\gamma} = {\sqrt{ɛ_{r} - {jx}} = {n = {\frac{1}{W} = {\frac{1}{W}{e^{{- j}\; \Psi}.}}}}}}}} & (31) \end{matrix}$

With the angle of incidence equal to the complex Brewster angle (θ_(i,B)), the Fresnel reflection coefficient of Equation (25) vanishes, as shown by

$\begin{matrix} {{\Gamma_{}\left( \theta_{i,B} \right)} = {\left. \frac{\sqrt{\left( {ɛ_{r} - {jx}} \right) - {\sin^{2}\theta_{i}}} - {\left( {ɛ_{r} - {jx}} \right)\cos \; \theta_{i}}}{\sqrt{\left( {ɛ_{r} - {jx}} \right) - {\sin^{2}\theta_{i}}} + {\left( {ɛ_{r} - {jx}} \right)\cos \; \theta_{i}}} \right|_{\theta_{i} = \theta_{i,B}} = 0.}} & (32) \end{matrix}$

By adjusting the complex field ratio of Equation (22), an incident field can be synthesized to be incident at a complex angle at which the reflection is reduced or eliminated. Establishing this ratio as n=√{square root over (ε_(r)−jx)} results in the synthesized electric field being incident at the complex Brewster angle, making the reflections vanish.

The concept of an electrical effective height can provide further insight into synthesizing an electric field with a complex angle of incidence with a guided surface waveguide probe 200. The electrical effective height (h_(eff)) has been defined as

$\begin{matrix} {h_{eff} = {\frac{1}{I_{0}}{\int_{0}^{h_{p}}{{I(z)}{dz}}}}} & (33) \end{matrix}$

for a monopole with a physical height (or length) of h_(p). Since the expression depends upon the magnitude and phase of the source distribution along the structure, the effective height (or length) is complex in general. The integration of the distributed current I(z) of the structure is performed over the physical height of the structure (h_(p)), and normalized to the ground current (I_(o)) flowing upward through the base (or input) of the structure. The distributed current along the structure can be expressed by

I(z)=I _(C) cos(β₀ z),  (34)

where β₀ is the propagation factor for current propagating on the structure. In the example of FIG. 3, I_(C) is the current that is distributed along the vertical structure of the guided surface waveguide probe 200 a.

For example, consider a feed network 209 that includes a low loss coil (e.g., a helical coil) at the bottom of the structure and a vertical feed line conductor connected between the coil and the charge terminal T₁. The phase delay due to the coil (or helical delay line) is θ_(c)=β_(p)l_(C), with a physical length of l_(C) and a propagation factor of

$\begin{matrix} {{\beta_{p} = {\frac{2\pi}{\lambda_{p}} = \frac{2\pi}{V_{f}\lambda_{0}}}},} & (35) \end{matrix}$

where V_(f) is the velocity factor on the structure, λ₀ is the wavelength at the supplied frequency, and λ_(p) is the propagation wavelength resulting from the velocity factor V_(f). The phase delay is measured relative to the ground (stake) current I₀.

In addition, the spatial phase delay along the length l_(w) of the vertical feed line conductor can be given by θ_(y)=β_(w)l_(w) where β_(w) is the propagation phase constant for the vertical feed line conductor. In some implementations, the spatial phase delay may be approximated by θ_(y)=β_(w)h_(p), since the difference between the physical height h_(p) of the guided surface waveguide probe 200 a and the vertical feed line conductor length l_(w) is much less than a wavelength at the supplied frequency (λ₀). As a result, the total phase delay through the coil and vertical feed line conductor is Φ=θ_(c)+θ_(y), and the current fed to the top of the coil from the bottom of the physical structure is

I _(C)(θ_(c)+θ_(y))=I ₀ e ^(jΦ),  (36)

with the total phase delay Φ measured relative to the ground (stake) current I₀. Consequently, the electrical effective height of a guided surface waveguide probe 200 can be approximated by

$\begin{matrix} {{h_{eff} = {{\frac{1}{I_{0}}{\int_{0}^{h_{p}}{I_{0}e^{j\; \Phi}{\cos \left( {\beta_{0}z} \right)}{dz}}}} \cong {h_{p}e^{j\; \Phi}}}},} & (37) \end{matrix}$

for the case where the physical height h_(p)<<λ₀. The complex effective height of a monopole, h_(eff)=h_(p) at an angle (or phase shift) of Φ, may be adjusted to cause the source fields to match a guided surface waveguide mode and cause a guided surface wave to be launched on the lossy conducting medium 203.

In the example of FIG. 5A, ray optics are used to illustrate the complex angle trigonometry of the incident electric field (E) having a complex Brewster angle of incidence (θ_(i,B)) at the Hankel crossover distance (R_(x)) 121. Recall from Equation (26) that, for a lossy conducting medium, the Brewster angle is complex and specified by

$\begin{matrix} {{\tan \; \theta_{i,B}} = {\sqrt{ɛ_{r} - {j\frac{\sigma}{{\omega ɛ}_{o}}}} = {n.}}} & (38) \end{matrix}$

Electrically, the geometric parameters are related by the electrical effective height (h_(eff)) of the charge terminal T₁ by

R _(x) tan ψ_(i,B) =R _(x) ×W=h _(eff) =h _(p) e ^(jΦ),  (39)

where ψ_(i,B)=(π/2)−θ_(i,B) is the Brewster angle measured from the surface of the lossy conducting medium. To couple into the guided surface waveguide mode, the wave tilt of the electric field at the Hankel crossover distance can be expressed as the ratio of the electrical effective height and the Hankel crossover distance

$\begin{matrix} {\frac{h_{eff}}{R_{x}} = {{\tan \; \psi_{i,B}} = {W_{Rx}.}}} & (40) \end{matrix}$

Since both the physical height (h_(p)) and the Hankel crossover distance (R_(x)) are real quantities, the angle (Ψ) of the desired guided surface wave tilt at the Hankel crossover distance (R_(x)) is equal to the phase (Φ) of the complex effective height (h_(eff)). This implies that by varying the phase at the supply point of the coil, and thus the phase shift in Equation (37), the phase, Φ, of the complex effective height can be manipulated to match the angle of the wave tilt, Ψ, of the guided surface waveguide mode at the Hankel crossover point 121: Φ=Ψ.

In FIG. 5A, a right triangle is depicted having an adjacent side of length R_(x) along the lossy conducting medium surface and a complex Brewster angle ψ_(i,B) measured between a ray 124 extending between the Hankel crossover point 121 at R_(x) and the center of the charge terminal T₁, and the lossy conducting medium surface 127 between the Hankel crossover point 121 and the charge terminal T₁. With the charge terminal T₁ positioned at physical height h_(p) and excited with a charge having the appropriate phase delay Φ, the resulting electric field is incident with the lossy conducting medium boundary interface at the Hankel crossover distance R_(x), and at the Brewster angle. Under these conditions, the guided surface waveguide mode can be excited without reflection or substantially negligible reflection.

If the physical height of the charge terminal T₁ is decreased without changing the phase shift Φ of the effective height (h_(eff)), the resulting electric field intersects the lossy conducting medium 203 at the Brewster angle at a reduced distance from the guided surface waveguide probe 200. FIG. 6 graphically illustrates the effect of decreasing the physical height of the charge terminal T₁ on the distance where the electric field is incident at the Brewster angle. As the height is decreased from h₃ through h₂ to h₁, the point where the electric field intersects with the lossy conducting medium (e.g., the Earth) at the Brewster angle moves closer to the charge terminal position. However, as Equation (39) indicates, the height H₁ (FIG. 3) of the charge terminal T₁ should be at or higher than the physical height (h_(p)) in order to excite the far-out component of the Hankel function. With the charge terminal T₁ positioned at or above the effective height (h_(eff)), the lossy conducting medium 203 can be illuminated at the Brewster angle of incidence (ψ_(i,B)=(π/2)−θ_(i,B)) at or beyond the Hankel crossover distance (R_(x)) 121 as illustrated in FIG. 5A. To reduce or minimize the bound charge on the charge terminal T₁, the height should be at least four times the spherical diameter (or equivalent spherical diameter) of the charge terminal T₁ as mentioned above.

A guided surface waveguide probe 200 can be configured to establish an electric field having a wave tilt that corresponds to a wave illuminating the surface of the lossy conducting medium 203 at a complex Brewster angle, thereby exciting radial surface currents by substantially mode-matching to a guided surface wave mode at (or beyond) the Hankel crossover point 121 at R_(x).

Referring to FIG. 7, shown is a graphical representation of an example of a guided surface waveguide probe 200 b that includes a charge terminal T₁. An AC source 212 acts as the excitation source for the charge terminal T₁, which is coupled to the guided surface waveguide probe 200 b through a feed network 209 (FIG. 3) comprising a coil 215 such as, e.g., a helical coil. In other implementations, the AC source 212 can be inductively coupled to the coil 215 through a primary coil. In some embodiments, an impedance matching network may be included to improve and/or maximize coupling of the AC source 212 to the coil 215.

As shown in FIG. 7, the guided surface waveguide probe 200 b can include the upper charge terminal T₁ (e.g., a sphere at height h_(p)) that is positioned along a vertical axis z that is substantially normal to the plane presented by the lossy conducting medium 203. A second medium 206 is located above the lossy conducting medium 203. The charge terminal T₁ has a self-capacitance C_(T). During operation, charge Q₁ is imposed on the terminal T₁ depending on the voltage applied to the terminal T₁ at any given instant.

In the example of FIG. 7, the coil 215 is coupled to a ground stake 218 at a first end and to the charge terminal T₁ via a vertical feed line conductor 221. In some implementations, the coil connection to the charge terminal T₁ can be adjusted using a tap 224 of the coil 215 as shown in FIG. 7. The coil 215 can be energized at an operating frequency by the AC source 212 through a tap 227 at a lower portion of the coil 215. In other implementations, the AC source 212 can be inductively coupled to the coil 215 through a primary coil.

The construction and adjustment of the guided surface waveguide probe 200 is based upon various operating conditions, such as the transmission frequency, conditions of the lossy conducting medium (e.g., soil conductivity a and relative permittivity ε_(r)), and size of the charge terminal T₁. The index of refraction can be calculated from Equations (10) and (11) as

n=√{square root over (ε_(r) −jx)},  (41)

where x=σ/ωε_(o) with ω=2πf. The conductivity a and relative permittivity ε_(r) can be determined through test measurements of the lossy conducting medium 203. The complex Brewster angle (θ_(i,B)) measured from the surface normal can also be determined from Equation (26) as

θ_(i,B)=arctan(√{square root over (ε_(r) −jx)}),  (42)

or measured from the surface as shown in FIG. 5A as

$\begin{matrix} {\psi_{i,B} = {\frac{\pi}{2} - {\theta_{i,B}.}}} & (43) \end{matrix}$

The wave tilt at the Hankel crossover distance (W_(Rx)) can also be found using Equation (40).

The Hankel crossover distance can also be found by equating the magnitudes of Equations (20b) and (21) for −jγρ, and solving for R_(x) as illustrated by FIG. 4. The electrical effective height can then be determined from Equation (39) using the Hankel crossover distance and the complex Brewster angle as

h _(eff) =h _(p) e ^(jΦ) =R _(x) tan ψ_(i,B).  (44)

As can be seen from Equation (44), the complex effective height (h_(eff)) includes a magnitude that is associated with the physical height (h_(p)) of the charge terminal T₁ and a phase delay (Φ) that is to be associated with the angle (Ψ) of the wave tilt at the Hankel crossover distance (R_(x)). With these variables and the selected charge terminal T₁ configuration, it is possible to determine the configuration of a guided surface waveguide probe 200.

With the charge terminal T₁ positioned at or above the physical height (h_(p)), the feed network 209 (FIG. 3) and/or the vertical feed line connecting the feed network to the charge terminal T₁ can be adjusted to match the phase (Φ) of the charge Q₁ on the charge terminal T₁ to the angle (Ψ) of the wave tilt (W). The size of the charge terminal T₁ can be chosen to provide a sufficiently large surface for the charge Q₁ imposed on the terminals. In general, it is desirable to make the charge terminal T₁ as large as practical. The size of the charge terminal T₁ should be large enough to avoid ionization of the surrounding air, which can result in electrical discharge or sparking around the charge terminal.

The phase delay θ_(c) of a helically-wound coil can be determined from Maxwell's equations as has been discussed by Corum, K. L. and J. F. Corum, “RF Coils, Helical Resonators and Voltage Magnification by Coherent Spatial Modes,” Microwave Review, Vol. 7, No. 2, September 2001, pp. 36-45, which is incorporated herein by reference in its entirety. For a helical coil with H/D>1, the ratio of the velocity of propagation (υ) of a wave along the coil's longitudinal axis to the speed of light (c), or the “velocity factor,” is given by

$\begin{matrix} {{V_{f} = {\frac{\upsilon}{c} = \frac{1}{\sqrt{1 + {20\left( \frac{D}{s} \right)^{2.5}\left( \frac{D}{\lambda_{o}} \right)^{0.5}}}}}},} & (45) \end{matrix}$

where H is the axial length of the solenoidal helix, D is the coil diameter, N is the number of turns of the coil, s=H/N is the turn-to-turn spacing (or helix pitch) of the coil, and λ_(o) is the free-space wavelength. Based upon this relationship, the electrical length, or phase delay, of the helical coil is given by

$\begin{matrix} {\theta_{c} = {{\beta_{p}H} = {{\frac{2\pi}{\lambda_{p}}H} = {\frac{2\pi}{V_{f}\lambda_{0}}{H.}}}}} & (46) \end{matrix}$

The principle is the same if the helix is wound spirally or is short and fat, but V_(f) and θ_(c) are easier to obtain by experimental measurement. The expression for the characteristic (wave) impedance of a helical transmission line has also been derived as

$\begin{matrix} {Z_{c} = {{\frac{60}{V_{f}}\left\lbrack {{\; {n\left( \frac{V_{f}\lambda_{0}}{D} \right)}} - 1.027} \right\rbrack}.}} & (47) \end{matrix}$

The spatial phase delay θ_(y) of the structure can be determined using the traveling wave phase delay of the vertical feed line conductor 221 (FIG. 7). The capacitance of a cylindrical vertical conductor above a prefect ground plane can be expressed as

$\begin{matrix} {{C_{A} = {\frac{2{\pi ɛ}_{o}h_{w}}{{\; {n\left( \frac{h}{a} \right)}} - 1}\mspace{14mu} {Farads}}},} & (48) \end{matrix}$

where h_(w) is the vertical length (or height) of the conductor and a is the radius (in mks units). As with the helical coil, the traveling wave phase delay of the vertical feed line conductor can be given by

$\begin{matrix} {{\theta_{y} = {{\beta_{w}h_{w}} = {{\frac{2\pi}{\lambda_{w}}h_{w}} = {\frac{2\pi}{V_{w}\lambda_{0}}h_{w}}}}},} & (49) \end{matrix}$

where β_(w) is the propagation phase constant for the vertical feed line conductor, h_(w) is the vertical length (or height) of the vertical feed line conductor, V_(w) is the velocity factor on the wire, λ₀ is the wavelength at the supplied frequency, and λ_(w) is the propagation wavelength resulting from the velocity factor V_(w). For a uniform cylindrical conductor, the velocity factor is a constant with V_(w)≈0.94, or in a range from about 0.93 to about 0.98. If the mast is considered to be a uniform transmission line, its average characteristic impedance can be approximated by

$\begin{matrix} {{Z_{w} = {\frac{60}{V_{w}}\left\lbrack {{\; {n\left( \frac{h_{w}}{a} \right)}} - 1} \right\rbrack}},} & (50) \end{matrix}$

where V_(w)≈0.94 for a uniform cylindrical conductor and a is the radius of the conductor. An alternative expression that has been employed in amateur radio literature for the characteristic impedance of a single-wire feed line can be given by

$\begin{matrix} {Z_{w} = {138\mspace{14mu} {{\log \left( \frac{1.123\; V_{w}\lambda_{0}}{2\pi \; a} \right)}.}}} & (51) \end{matrix}$

Equation (51) implies that Z_(w) for a single-wire feeder varies with frequency. The phase delay can be determined based upon the capacitance and characteristic impedance.

With a charge terminal T₁ positioned over the lossy conducting medium 203 as shown in FIG. 3, the feed network 209 can be adjusted to excite the charge terminal T₁ with the phase shift (Φ) of the complex effective height (h_(eff)) equal to the angle (Ψ) of the wave tilt at the Hankel crossover distance, or Φ=Ψ. When this condition is met, the electric field produced by the charge oscillating Q₁ on the charge terminal T₁ is coupled into a guided surface waveguide mode traveling along the surface of a lossy conducting medium 203. For example, if the Brewster angle (θ_(i,B)), the phase delay (θ_(y)) associated with the vertical feed line conductor 221 (FIG. 7), and the configuration of the coil 215 (FIG. 7) are known, then the position of the tap 224 (FIG. 7) can be determined and adjusted to impose an oscillating charge Q₁ on the charge terminal T₁ with phase Φ=Ψ. The position of the tap 224 may be adjusted to maximize coupling the traveling surface waves into the guided surface waveguide mode. Excess coil length beyond the position of the tap 224 can be removed to reduce the capacitive effects. The vertical wire height and/or the geometrical parameters of the helical coil may also be varied.

The coupling to the guided surface waveguide mode on the surface of the lossy conducting medium 203 can be improved and/or optimized by tuning the guided surface waveguide probe 200 for standing wave resonance with respect to a complex image plane associated with the charge Q₁ on the charge terminal T₁. By doing this, the performance of the guided surface waveguide probe 200 can be adjusted for increased and/or maximum voltage (and thus charge Q₁) on the charge terminal T₁. Referring back to FIG. 3, the effect of the lossy conducting medium 203 in Region 1 can be examined using image theory analysis.

Physically, an elevated charge Q₁ placed over a perfectly conducting plane attracts the free charge on the perfectly conducting plane, which then “piles up” in the region under the elevated charge Q₁. The resulting distribution of “bound” electricity on the perfectly conducting plane is similar to a bell-shaped curve. The superposition of the potential of the elevated charge Q₁, plus the potential of the induced “piled up” charge beneath it, forces a zero equipotential surface for the perfectly conducting plane. The boundary value problem solution that describes the fields in the region above the perfectly conducting plane may be obtained using the classical notion of image charges, where the field from the elevated charge is superimposed with the field from a corresponding “image” charge below the perfectly conducting plane.

This analysis may also be used with respect to a lossy conducting medium 203 by assuming the presence of an effective image charge Q₁′ beneath the guided surface waveguide probe 200. The effective image charge Q₁′ coincides with the charge Q₁ on the charge terminal T₁ about a conducting image ground plane 130, as illustrated in FIG. 3. However, the image charge Q₁′ is not merely located at some real depth and 180° out of phase with the primary source charge Q₁ on the charge terminal T₁, as they would be in the case of a perfect conductor. Rather, the lossy conducting medium 203 (e.g., a terrestrial medium) presents a phase shifted image. That is to say, the image charge Q₁′ is at a complex depth below the surface (or physical boundary) of the lossy conducting medium 203. For a discussion of complex image depth, reference is made to Wait, J. R., “Complex Image Theory—Revisited,” IEEE Antennas and Propagation Magazine, Vol. 33, No. 4, August 1991, pp. 27-29, which is incorporated herein by reference in its entirety.

Instead of the image charge Q₁′ being at a depth that is equal to the physical height (H₁) of the charge Q₁, the conducting image ground plane 130 (representing a perfect conductor) is located at a complex depth of z=−d/2 and the image charge Q₁′ appears at a complex depth (i.e., the “depth” has both magnitude and phase), given by −D₁=−(d/2+d/2+H₁)≠H₁. For vertically polarized sources over the Earth,

$\begin{matrix} {{d = {{\frac{\sqrt[2]{\gamma_{e}^{2} + k_{0}^{2}}}{\gamma_{e}^{2}} \approx \frac{2}{\gamma_{e}}} = {{d_{r} + {jd}_{i}} = {{d}{\angle\zeta}}}}},} & (52) \end{matrix}$

where

γ_(e) ² =jωμ ₁σ₁−ω²μ₁ε₁, and  (53)

k _(o)=ω√{square root over (μ_(o)ε_(o))},  (54)

as indicated in Equation (12). The complex spacing of the image charge, in turn, implies that the external field will experience extra phase shifts not encountered when the interface is either a dielectric or a perfect conductor. In the lossy conducting medium, the wave front normal is parallel to the tangent of the conducting image ground plane 130 at z=−d/2, and not at the boundary interface between Regions 1 and 2.

Consider the case illustrated in FIG. 8A where the lossy conducting medium 203 is a finitely conducting Earth 133 with a physical boundary 136. The finitely conducting Earth 133 may be replaced by a perfectly conducting image ground plane 139 as shown in FIG. 8B, which is located at a complex depth z₁ below the physical boundary 136. This equivalent representation exhibits the same impedance when looking down into the interface at the physical boundary 136. The equivalent representation of FIG. 8B can be modeled as an equivalent transmission line, as shown in FIG. 8C. The cross-section of the equivalent structure is represented as a (z-directed) end-loaded transmission line, with the impedance of the perfectly conducting image plane being a short circuit (z_(s)=0). The depth z₁ can be determined by equating the TEM wave impedance looking down at the Earth to an image ground plane impedance z_(in) seen looking into the transmission line of FIG. 8C.

In the case of FIG. 8A, the propagation constant and wave intrinsic impedance in the upper region (air) 142 are

$\begin{matrix} {{\gamma_{o} = {{j\; \omega \sqrt{\mu_{o}ɛ_{o}}} = {0 + {j\; \beta_{o}}}}},{and}} & (55) \\ {z_{o} = {\frac{j\; {\omega\mu}_{o}}{\gamma_{o}} = {\sqrt{\frac{\mu_{o}}{ɛ_{o}}}.}}} & (56) \end{matrix}$

In the lossy Earth 133, the propagation constant and wave intrinsic impedance are

$\begin{matrix} {{\gamma_{e} = \sqrt{j\; {{\omega\mu}_{1}\left( {\sigma_{1} + {j\; {\omega ɛ}_{1}}} \right)}}},{and}} & (57) \\ {Z_{e} = {\frac{j\; {\omega\mu}_{1}}{\gamma_{e}}.}} & (58) \end{matrix}$

For normal incidence, the equivalent representation of FIG. 8B is equivalent to a TEM transmission line whose characteristic impedance is that of air (z_(o)), with propagation constant of γ_(o), and whose length is z₁. As such, the image ground plane impedance Z_(in) seen at the interface for the shorted transmission line of FIG. 8C is given by

Z _(in) =Z _(o) tan h(γ_(o) z ₁).  (59)

Equating the image ground plane impedance Z_(in) associated with the equivalent model of FIG. 8C to the normal incidence wave impedance of FIG. 8A and solving for z₁ gives the distance to a short circuit (the perfectly conducting image ground plane 139) as

$\begin{matrix} {{z_{1} = {{\frac{1}{\gamma_{o}}{\tanh^{- 1}\left( \frac{Z_{e}}{Z_{o}} \right)}} = {{\frac{1}{\gamma_{o}}{\tanh^{- 1}\left( \frac{\gamma_{o}}{\gamma_{e}} \right)}} \approx \frac{1}{\gamma_{e}}}}},} & (60) \end{matrix}$

where only the first term of the series expansion for the inverse hyperbolic tangent is considered for this approximation. Note that in the air region 142, the propagation constant is γ_(o)=jβ_(o), so Z_(in)=jZ_(o) tan β_(o)z₁ (which is a purely imaginary quantity for a real z₁), but z_(e) is a complex value if σ≠0. Therefore, Z_(in)=Z_(e) only when z₁ is a complex distance.

Since the equivalent representation of FIG. 8B includes a perfectly conducting image ground plane 139, the image depth for a charge or current lying at the surface of the Earth (physical boundary 136) is equal to distance z₁ on the other side of the image ground plane 139, or d=2×z₁ beneath the Earth's surface (which is located at z=0). Thus, the distance to the perfectly conducting image ground plane 139 can be approximated by

$\begin{matrix} {d = {{2z_{1}} \approx {\frac{2}{\gamma_{e}}.}}} & (61) \end{matrix}$

Additionally, the “image charge” will be “equal and opposite” to the real charge, so the potential of the perfectly conducting image ground plane 139 at depth z₁=−d/2 will be zero.

If a charge Q₁ is elevated a distance H₁ above the surface of the Earth as illustrated in FIG. 3, then the image charge Q₁′ resides at a complex distance of D₁=d+H₁ below the surface, or a complex distance of d/2+H₁ below the image ground plane 130. The guided surface waveguide probe 200 b of FIG. 7 can be modeled as an equivalent single-wire transmission line image plane model that can be based upon the perfectly conducting image ground plane 139 of FIG. 8B. FIG. 9A shows an example of the equivalent single-wire transmission line image plane model, and FIG. 9B illustrates an example of the equivalent classic transmission line model, including the shorted transmission line of FIG. 8C.

In the equivalent image plane models of FIGS. 9A and 9B, Φ=θ_(y)+θ_(c) is the traveling wave phase delay of the guided surface waveguide probe 200 referenced to Earth 133 (or the lossy conducting medium 203), θ_(c)=β_(p)H is the electrical length of the coil 215 (FIG. 7), of physical length H, expressed in degrees, θ_(y)=β_(w)h_(w) is the electrical length of the vertical feed line conductor 221 (FIG. 7), of physical length h_(w), expressed in degrees, and θ_(d)=β_(o) d/2 is the phase shift between the image ground plane 139 and the physical boundary 136 of the Earth 133 (or lossy conducting medium 203). In the example of FIGS. 9A and 9B, Z_(w) is the characteristic impedance of the elevated vertical feed line conductor 221 in ohms, Z_(c) is the characteristic impedance of the coil 215 in ohms, and Z_(O) is the characteristic impedance of free space.

At the base of the guided surface waveguide probe 200, the impedance seen “looking up” into the structure is Z_(↑)=Z_(base). With a load impedance of:

$\begin{matrix} {{Z_{L} = \frac{1}{j\; \omega \; C_{T}}},} & (62) \end{matrix}$

where C_(T) is the self-capacitance of the charge terminal T₁, the impedance seen “looking up” into the vertical feed line conductor 221 (FIG. 7) is given by:

$\begin{matrix} {{Z_{2} = {{Z_{W}\frac{Z_{L} + {Z_{w}{\tanh \left( {j\; \beta_{w}h_{w}} \right)}}}{Z_{w} + {Z_{L}{\tanh \left( {j\; \beta_{w}h_{w}} \right)}}}} = {Z_{W}\frac{Z_{L} + {Z_{w}{\tanh \left( {j\; \theta_{y}} \right)}}}{Z_{w} + {Z_{L}{\tanh \left( {j\; \theta_{y}} \right)}}}}}},} & (63) \end{matrix}$

and the impedance seen “looking up” into the coil 215 (FIG. 7) is given by:

$\begin{matrix} {Z_{base} = {{Z_{c}\frac{Z_{2} + {Z_{c}{\tanh \left( {j\; \beta_{p}H} \right)}}}{Z_{c} + {Z_{2}{\tanh \left( {j\; \beta_{p}H} \right)}}}} = {Z_{c}{\frac{Z_{2} + {Z_{c}{\tanh \left( {j\; \theta_{c}} \right)}}}{Z_{c} + {Z_{2}{\tanh \left( {j\; \theta_{c}} \right)}}}.}}}} & (64) \end{matrix}$

At the base of the guided surface waveguide probe 200, the impedance seen “looking down” into the lossy conducting medium 203 is Z_(↓)=Z_(in), which is given by:

$\begin{matrix} {{Z_{in} = {{Z_{o}\frac{Z_{s} + {Z_{o}{\tanh \left\lbrack {j\; {\beta_{o}\left( {d/2} \right)}} \right\rbrack}}}{Z_{o} + {Z_{s}{\tanh \left\lbrack {j\; {\beta_{o}\left( {d/2} \right)}} \right\rbrack}}}} = {Z_{o}{\tanh \left( {j\; \theta_{d}} \right)}}}},} & (65) \end{matrix}$

where Z_(s)=0.

Neglecting losses, the equivalent image plane model can be tuned to resonance when Z_(↓)+Z_(↑)=0 at the physical boundary 136. Or, in the low loss case, X_(↓)+X_(↑)=0 at the physical boundary 136, where X is the corresponding reactive component. Thus, the impedance at the physical boundary 136 “looking up” into the guided surface waveguide probe 200 is the conjugate of the impedance at the physical boundary 136 “looking down” into the lossy conducting medium 203. By adjusting the load impedance Z_(L) of the charge terminal T₁ while maintaining the traveling wave phase delay Φ equal to the angle of the media's wave tilt Ψ, so that Φ=Ψ, which improves and/or maximizes coupling of the probe's electric field to a guided surface waveguide mode along the surface of the lossy conducting medium 203 (e.g., Earth), the equivalent image plane models of FIGS. 9A and 9B can be tuned to resonance with respect to the image ground plane 139. In this way, the impedance of the equivalent complex image plane model is purely resistive, which maintains a superposed standing wave on the probe structure that maximizes the voltage and elevated charge on terminal T₁, and by equations (1)-(3) and (16) maximizes the propagating surface wave.

It follows from the Hankel solutions, that the guided surface wave excited by the guided surface waveguide probe 200 is an outward propagating traveling wave. The source distribution along the feed network 209 between the charge terminal T₁ and the ground stake 218 of the guided surface waveguide probe 200 (FIGS. 3 and 7) is actually composed of a superposition of a traveling wave plus a standing wave on the structure. With the charge terminal T₁ positioned at or above the physical height h_(p), the phase delay of the traveling wave moving through the feed network 209 is matched to the angle of the wave tilt associated with the lossy conducting medium 203. This mode-matching allows the traveling wave to be launched along the lossy conducting medium 203. Once the phase delay has been established for the traveling wave, the load impedance Z_(L) of the charge terminal T₁ is adjusted to bring the probe structure into standing wave resonance with respect to the image ground plane (130 of FIG. 3 or 139 of FIG. 8), which is at a complex depth of −d/2. In that case, the impedance seen from the image ground plane has zero reactance and the charge on the charge terminal T₁ is maximized.

The distinction between the traveling wave phenomenon and standing wave phenomena is that (1) the phase delay of traveling waves (θ=βd) on a section of transmission line of length d (sometimes called a “delay line”) is due to propagation time delays; whereas (2) the position-dependent phase of standing waves (which are composed of forward and backward propagating waves) depends on both the line length propagation time delay and impedance transitions at interfaces between line sections of different characteristic impedances. In addition to the phase delay that arises due to the physical length of a section of transmission line operating in sinusoidal steady-state, there is an extra reflection coefficient phase at impedance discontinuities that is due to the ratio of Z_(oa)/Z_(ob), where Z_(oa) and Z_(ob) are the characteristic impedances of two sections of a transmission line such as, e.g., a helical coil section of characteristic impedance Z_(oa)=Z_(c) (FIG. 9B) and a straight section of vertical feed line conductor of characteristic impedance Z_(ob)=Z_(w) (FIG. 9B).

As a result of this phenomenon, two relatively short transmission line sections of widely differing characteristic impedance may be used to provide a very large phase shift. For example, a probe structure composed of two sections of transmission line, one of low impedance and one of high impedance, together totaling a physical length of, say, 0.05λ, may be fabricated to provide a phase shift of 90° which is equivalent to a 0.25λ resonance. This is due to the large jump in characteristic impedances. In this way, a physically short probe structure can be electrically longer than the two physical lengths combined. This is illustrated in FIGS. 9A and 9B, where the discontinuities in the impedance ratios provide large jumps in phase. The impedance discontinuity provides a substantial phase shift where the sections are joined together.

Referring to FIG. 10, shown is a flow chart 150 illustrating an example of adjusting a guided surface waveguide probe 200 (FIGS. 3 and 7) to substantially mode-match to a guided surface waveguide mode on the surface of the lossy conducting medium, which launches a guided surface traveling wave along the surface of a lossy conducting medium 203 (FIG. 3). Beginning with 153, the charge terminal T₁ of the guided surface waveguide probe 200 is positioned at a defined height above a lossy conducting medium 203. Utilizing the characteristics of the lossy conducting medium 203 and the operating frequency of the guided surface waveguide probe 200, the Hankel crossover distance can also be found by equating the magnitudes of Equations (20b) and (21) for −jγρ, and solving for R_(x) as illustrated by FIG. 4. The complex index of refraction (n) can be determined using Equation (41), and the complex Brewster angle (θ_(i,B)) can then be determined from Equation (42). The physical height (h_(p)) of the charge terminal T₁ can then be determined from Equation (44). The charge terminal T₁ should be at or higher than the physical height (h_(p)) in order to excite the far-out component of the Hankel function. This height relationship is initially considered when launching surface waves. To reduce or minimize the bound charge on the charge terminal T₁, the height should be at least four times the spherical diameter (or equivalent spherical diameter) of the charge terminal T₁.

At 156, the electrical phase delay Φ of the elevated charge Q₁ on the charge terminal T₁ is matched to the complex wave tilt angle Ψ. The phase delay (θ_(c)) of the helical coil and/or the phase delay (θ_(y)) of the vertical feed line conductor can be adjusted to make Φ equal to the angle (Ψ) of the wave tilt (W). Based on Equation (31), the angle (Ψ) of the wave tilt can be determined from:

$\begin{matrix} {W = {\frac{E_{\rho}}{E_{z}} = {\frac{1}{\tan \; \theta_{i,B}} = {\frac{1}{n} = {{W}{e^{j\; \Psi}.}}}}}} & (66) \end{matrix}$

The electrical phase Φ can then be matched to the angle of the wave tilt. This angular (or phase) relationship is next considered when launching surface waves. For example, the electrical phase delay Φ=θ_(c)+θ_(y) can be adjusted by varying the geometrical parameters of the coil 215 (FIG. 7) and/or the length (or height) of the vertical feed line conductor 221 (FIG. 7). By matching Φ=Ψ, an electric field can be established at or beyond the Hankel crossover distance (R_(x)) with a complex Brewster angle at the boundary interface to excite the surface waveguide mode and launch a traveling wave along the lossy conducting medium 203.

Next at 159, the load impedance of the charge terminal T₁ is tuned to resonate the equivalent image plane model of the guided surface waveguide probe 200. The depth (d/2) of the conducting image ground plane 139 of FIGS. 9A and 9B (or 130 of FIG. 3) can be determined using Equations (52), (53) and (54) and the values of the lossy conducting medium 203 (e.g., the Earth), which can be measured. Using that depth, the phase shift (θ_(d)) between the image ground plane 139 and the physical boundary 136 of the lossy conducting medium 203 can be determined using θ_(d)=β_(o) d/2. The impedance (Z_(in)) as seen “looking down” into the lossy conducting medium 203 can then be determined using Equation (65). This resonance relationship can be considered to maximize the launched surface waves.

Based upon the adjusted parameters of the coil 215 and the length of the vertical feed line conductor 221, the velocity factor, phase delay, and impedance of the coil 215 and vertical feed line conductor 221 can be determined using Equations (45) through (51). In addition, the self-capacitance (C_(T)) of the charge terminal T₁ can be determined using, e.g., Equation (24). The propagation factor (β_(p)) of the coil 215 can be determined using Equation (35) and the propagation phase constant (β_(w)) for the vertical feed line conductor 221 can be determined using Equation (49). Using the self-capacitance and the determined values of the coil 215 and vertical feed line conductor 221, the impedance (Z_(base)) of the guided surface waveguide probe 200 as seen “looking up” into the coil 215 can be determined using Equations (62), (63) and (64).

The equivalent image plane model of the guided surface waveguide probe 200 can be tuned to resonance by adjusting the load impedance Z_(L) such that the reactance component X_(base) of Z_(base) cancels out the reactance component X_(in) of Z_(in), or X_(base)+X_(in)=0. Thus, the impedance at the physical boundary 136 “looking up” into the guided surface waveguide probe 200 is the conjugate of the impedance at the physical boundary 136 “looking down” into the lossy conducting medium 203. The load impedance Z_(L) can be adjusted by varying the capacitance (C_(T)) of the charge terminal T₁ without changing the electrical phase delay Φ=θ_(c)+θ_(y) of the charge terminal T₁. An iterative approach may be taken to tune the load impedance Z_(L) for resonance of the equivalent image plane model with respect to the conducting image ground plane 139 (or 130). In this way, the coupling of the electric field to a guided surface waveguide mode along the surface of the lossy conducting medium 203 (e.g., Earth) can be improved and/or maximized.

This may be better understood by illustrating the situation with a numerical example. Consider a guided surface waveguide probe 200 comprising a top-loaded vertical stub of physical height h_(p) with a charge terminal T₁ at the top, where the charge terminal T₁ is excited through a helical coil and vertical feed line conductor at an operational frequency (f_(o)) of 1.85 MHz. With a height (H₁) of 16 feet and the lossy conducting medium 203 (e.g., Earth) having a relative permittivity of ε_(r)=15 and a conductivity of σ₁=0.010 mhos/m, several surface wave propagation parameters can be calculated for f_(o)=1.850 MHz. Under these conditions, the Hankel crossover distance can be found to be R_(x)=54.5 feet with a physical height of h_(p)=5.5 feet, which is well below the actual height of the charge terminal T₁. While a charge terminal height of H₁=5.5 feet could have been used, the taller probe structure reduced the bound capacitance, permitting a greater percentage of free charge on the charge terminal T₁ providing greater field strength and excitation of the traveling wave.

The wave length can be determined as:

$\begin{matrix} {{\lambda_{o} = {\frac{c}{f_{o}} = {162.162\mspace{14mu} {meters}}}},} & (67) \end{matrix}$

where c is the speed of light. The complex index of refraction is:

n=√{square root over (ε_(r) −jx)}=7.529−j6.546,  (68)

from Equation (41), where x=σ₁/ωε_(o) with ω=2πf_(o), and the complex Brewster angle is:

θ_(i,B)=arctan(√{square root over (ε_(r) −jx)})=85.6−j3.744°.  (69)

from Equation (42). Using Equation (66), the wave tilt values can be determined to be:

$\begin{matrix} {W = {\frac{1}{\tan \; \theta_{i,B}} = {\frac{1}{n} = {{{W}e^{j\; \Psi}} = {0.101{e^{j\; 40.614{^\circ}}.}}}}}} & (70) \end{matrix}$

Thus, the helical coil can be adjusted to match Φ=Ψ=40.614°

The velocity factor of the vertical feed line conductor (approximated as a uniform cylindrical conductor with a diameter of 0.27 inches) can be given as V_(w)≈0.93. Since h_(p)<<λ_(o), the propagation phase constant for the vertical feed line conductor can be approximated as:

$\begin{matrix} {\beta_{w} = {\frac{2\pi}{\lambda_{w}} = {\frac{2\pi}{V_{w}\lambda_{0}} = {0.042\mspace{14mu} {m^{- 1}.}}}}} & (71) \end{matrix}$

From Equation (49) the phase delay of the vertical feed line conductor is:

θ_(y)=β_(w) h _(w)≈β_(w) h _(p)=11.640°.  (72)

By adjusting the phase delay of the helical coil so that θ_(c)=28.974°=40.614°−11.640°, Φ will equal Ψ to match the guided surface waveguide mode. To illustrate the relationship between Φ and Ψ, FIG. 11 shows a plot of both over a range of frequencies. As both Φ and Ψ are frequency dependent, it can be seen that their respective curves cross over each other at approximately 1.85 MHz.

For a helical coil having a conductor diameter of 0.0881 inches, a coil diameter (D) of 30 inches and a turn-to-turn spacing (s) of 4 inches, the velocity factor for the coil can be determined using Equation (45) as:

$\begin{matrix} {{V_{f} = {\frac{1}{\sqrt{1 + {20\left( \frac{D}{s} \right)^{2.5}\left( \frac{D}{\lambda_{o}} \right)^{0.5}}}} = 0.069}},} & (73) \end{matrix}$

and the propagation factor from Equation (35) is:

$\begin{matrix} {\beta_{p} = {\frac{2\pi}{V_{f}\lambda_{0}} = {0.564\mspace{14mu} {m^{- 1}.}}}} & (74) \end{matrix}$

With θ_(c)=28.974°, the axial length of the solenoidal helix (H) can be determined using Equation (46) such that:

$\begin{matrix} {H = {\frac{\theta_{c}}{\beta_{p}} = {35.2732\mspace{14mu} {{inches}.}}}} & (75) \end{matrix}$

This height determines the location on the helical coil where the vertical feed line conductor is connected, resulting in a coil with 8.818 turns (N=H/s).

With the traveling wave phase delay of the coil and vertical feed line conductor adjusted to match the wave tilt angle (Φ=θ_(c)+θ_(y)=Ψ), the load impedance (Z_(L)) of the charge terminal T₁ can be adjusted for standing wave resonance of the equivalent image plane model of the guided surface wave probe 200. From the measured permittivity, conductivity and permeability of the Earth, the radial propagation constant can be determined using Equation (57)

γ_(e)=√{square root over (jωu ₁(σ₁ +jωε ₁))}=0.25+j0.292m ⁻¹,  (76)

And the complex depth of the conducting image ground plane can be approximated from Equation (52) as:

$\begin{matrix} {{{d \approx \frac{2}{\gamma_{e}}} = {3.364 + {j\mspace{14mu} 3.963\mspace{14mu} {meters}}}},} & (77) \end{matrix}$

with a corresponding phase shift between the conducting image ground plane and the physical boundary of the Earth given by:

θ_(d)=β_(o)(d/2)=4.015−j4.73°.  (78)

Using Equation (65), the impedance seen “looking down” into the lossy conducting medium 203 (i.e., Earth) can be determined as:

Z _(in) =Z _(o) tan h(jθ _(d))=R _(in) +jX _(in)=31.191+j26.27 ohms.  (79)

By matching the reactive component (X_(in)) seen “looking down” into the lossy conducting medium 203 with the reactive component (X_(base)) seen “looking up” into the guided surface wave probe 200, the coupling into the guided surface waveguide mode may be maximized. This can be accomplished by adjusting the capacitance of the charge terminal T₁ without changing the traveling wave phase delays of the coil and vertical feed line conductor. For example, by adjusting the charge terminal capacitance (C_(T)) to 61.8126 pF, the load impedance from Equation (62) is:

$\begin{matrix} {{Z_{L} = {\frac{1}{j\; \omega \; C_{T}} = {{- j}\mspace{14mu} 1392\mspace{14mu} {ohms}}}},} & (80) \end{matrix}$

and the reactive components at the boundary are matched.

Using Equation (51), the impedance of the vertical feed line conductor (having a diameter (2a) of 0.27 inches) is given as

$\begin{matrix} {{Z_{w} = {{138{\log \left( \frac{1.123V_{w}\lambda_{0}}{2\pi \; a} \right)}} = {537.534\mspace{14mu} {ohms}}}},} & (81) \end{matrix}$

and the impedance seen “looking up” into the vertical feed line conductor is given by Equation (63) as:

$\begin{matrix} {Z_{2} = {{Z_{W}\frac{Z_{L} + {Z_{w}{\tanh \left( {j\; \theta_{y}} \right)}}}{Z_{w} + {Z_{L}{\tanh \left( {j\; \theta_{y}} \right)}}}} = {{- j}\mspace{11mu} 835.438\mspace{14mu} {{ohms}.}}}} & (82) \end{matrix}$

Using Equation (47), the characteristic impedance of the helical coil is given as

$\begin{matrix} {{Z_{c} = {{\frac{60}{V_{f}}\left\lbrack {{\; {n\left( \frac{V_{f}\lambda_{0}}{D} \right)}} - 1.027} \right\rbrack} = {1446\mspace{14mu} {ohms}}}},} & (83) \end{matrix}$

and the impedance seen “looking up” into the coil at the base is given by Equation (64) as:

$\begin{matrix} {Z_{base} = {{Z_{c}\frac{Z_{2} + {Z_{c}{\tanh \left( {j\; \theta_{c}} \right)}}}{Z_{c} + {Z_{2}{\tanh \left( {j\; \theta_{c}} \right)}}}} = {{- j}\mspace{11mu} 26.271\mspace{14mu} {{ohms}.}}}} & (84) \end{matrix}$

When compared to the solution of Equation (79), it can be seen that the reactive components are opposite and approximately equal, and thus are conjugates of each other. Thus, the impedance (Z_(ip)) seen “looking up” into the equivalent image plane model of FIGS. 9A and 9B from the perfectly conducting image ground plane is only resistive or Z_(ip)=R+j0.

When the electric fields produced by a guided surface waveguide probe 200 (FIG. 3) are established by matching the traveling wave phase delay of the feed network to the wave tilt angle and the probe structure is resonated with respect to the perfectly conducting image ground plane at complex depth z=−d/2, the fields are substantially mode-matched to a guided surface waveguide mode on the surface of the lossy conducting medium, a guided surface traveling wave is launched along the surface of the lossy conducting medium. As illustrated in FIG. 1, the guided field strength curve 103 of the guided electromagnetic field has a characteristic exponential decay of e^(−αd)/√{square root over (d)} exhibits a distinctive knee 109 on the log-log scale.

In summary, both analytically and experimentally, the traveling wave component on the structure of the guided surface waveguide probe 200 has a phase delay (Φ) at its upper terminal that matches the angle (Ψ) of the wave tilt of the surface traveling wave (Φ=Ψ). Under this condition, the surface waveguide may be considered to be “mode-matched”. Furthermore, the resonant standing wave component on the structure of the guided surface waveguide probe 200 has a V_(MAX) at the charge terminal T₁ and a V_(MIN) down at the image plane 139 (FIG. 8B) where Z_(ip)=R_(ip)+j 0 at a complex depth of z=−d/2, not at the connection at the physical boundary 136 of the lossy conducting medium 203 (FIG. 8B). Lastly, the charge terminal T₁ is of sufficient height H₁ of FIG. 3 (h≥R_(x) tan ψ_(i,B)) so that electromagnetic waves incident onto the lossy conducting medium 203 at the complex Brewster angle do so out at a distance (≥R_(x)) where the 1/√{square root over (r)} term is predominant. Receive circuits can be utilized with one or more guided surface waveguide probes to facilitate wireless transmission and/or power delivery systems.

Referring back to FIG. 3, operation of a guided surface waveguide probe 200 may be controlled to adjust for variations in operational conditions associated with the guided surface waveguide probe 200. For example, an adaptive probe control system 230 can be used to control the feed network 209 and/or the charge terminal T₁ to control the operation of the guided surface waveguide probe 200. Operational conditions can include, but are not limited to, variations in the characteristics of the lossy conducting medium 203 (e.g., conductivity a and relative permittivity ε_(r)), variations in field strength and/or variations in loading of the guided surface waveguide probe 200. As can be seen from Equations (31), (41) and (42), the index of refraction (n), the complex Brewster angle (θ_(i,B)), and the wave tilt (|W|e^(jΨ)) can be affected by changes in soil conductivity and permittivity resulting from, e.g., weather conditions.

Equipment such as, e.g., conductivity measurement probes, permittivity sensors, ground parameter meters, field meters, current monitors and/or load receivers can be used to monitor for changes in the operational conditions and provide information about current operational conditions to the adaptive probe control system 230. The probe control system 230 can then make one or more adjustments to the guided surface waveguide probe 200 to maintain specified operational conditions for the guided surface waveguide probe 200. For instance, as the moisture and temperature vary, the conductivity of the soil will also vary. Conductivity measurement probes and/or permittivity sensors may be located at multiple locations around the guided surface waveguide probe 200. Generally, it would be desirable to monitor the conductivity and/or permittivity at or about the Hankel crossover distance R_(x) for the operational frequency. Conductivity measurement probes and/or permittivity sensors may be located at multiple locations (e.g., in each quadrant) around the guided surface waveguide probe 200.

The conductivity measurement probes and/or permittivity sensors can be configured to evaluate the conductivity and/or permittivity on a periodic basis and communicate the information to the probe control system 230. The information may be communicated to the probe control system 230 through a network such as, but not limited to, a LAN, WLAN, cellular network, or other appropriate wired or wireless communication network. Based upon the monitored conductivity and/or permittivity, the probe control system 230 may evaluate the variation in the index of refraction (n), the complex Brewster angle (θ_(i,B)), and/or the wave tilt (|W|e^(jΨ)) and adjust the guided surface waveguide probe 200 to maintain the phase delay (Φ) of the feed network 209 equal to the wave tilt angle (Ψ) and/or maintain resonance of the equivalent image plane model of the guided surface waveguide probe 200. This can be accomplished by adjusting, e.g., θ_(y), θ_(c) and/or C_(T). For instance, the probe control system 230 can adjust the self-capacitance of the charge terminal T₁ and/or the phase delay (θ_(y), θ_(c)) applied to the charge terminal T₁ to maintain the electrical launching efficiency of the guided surface wave at or near its maximum. For example, the self-capacitance of the charge terminal T₁ can be varied by changing the size of the terminal. The charge distribution can also be improved by increasing the size of the charge terminal T₁, which can reduce the chance of an electrical discharge from the charge terminal T₁. In other embodiments, the charge terminal T₁ can include a variable inductance that can be adjusted to change the load impedance Z_(L). The phase applied to the charge terminal T₁ can be adjusted by varying the tap position on the coil 215 (FIG. 7), and/or by including a plurality of predefined taps along the coil 215 and switching between the different predefined tap locations to maximize the launching efficiency.

Field or field strength (FS) meters may also be distributed about the guided surface waveguide probe 200 to measure field strength of fields associated with the guided surface wave. The field or FS meters can be configured to detect the field strength and/or changes in the field strength (e.g., electric field strength) and communicate that information to the probe control system 230. The information may be communicated to the probe control system 230 through a network such as, but not limited to, a LAN, WLAN, cellular network, or other appropriate communication network. As the load and/or environmental conditions change or vary during operation, the guided surface waveguide probe 200 may be adjusted to maintain specified field strength(s) at the FS meter locations to ensure appropriate power transmission to the receivers and the loads they supply.

For example, the phase delay (Φ=θ_(y)+θ_(c)) applied to the charge terminal T₁ can be adjusted to match the wave tilt angle (Ψ). By adjusting one or both phase delays, the guided surface waveguide probe 200 can be adjusted to ensure the wave tilt corresponds to the complex Brewster angle. This can be accomplished by adjusting a tap position on the coil 215 (FIG. 7) to change the phase delay supplied to the charge terminal T₁. The voltage level supplied to the charge terminal T₁ can also be increased or decreased to adjust the electric field strength. This may be accomplished by adjusting the output voltage of the excitation source 212 or by adjusting or reconfiguring the feed network 209. For instance, the position of the tap 227 (FIG. 7) for the AC source 212 can be adjusted to increase the voltage seen by the charge terminal T₁. Maintaining field strength levels within predefined ranges can improve coupling by the receivers, reduce ground current losses, and avoid interference with transmissions from other guided surface waveguide probes 200.

The probe control system 230 can be implemented with hardware, firmware, software executed by hardware, or a combination thereof. For example, the probe control system 230 can include processing circuitry including a processor and a memory, both of which can be coupled to a local interface such as, for example, a data bus with an accompanying control/address bus as can be appreciated by those with ordinary skill in the art. A probe control application may be executed by the processor to adjust the operation of the guided surface waveguide probe 200 based upon monitored conditions. The probe control system 230 can also include one or more network interfaces for communicating with the various monitoring devices. Communications can be through a network such as, but not limited to, a LAN, WLAN, cellular network, or other appropriate communication network. The probe control system 230 may comprise, for example, a computer system such as a server, desktop computer, laptop, or other system with like capability.

Referring back to the example of FIG. 5A, the complex angle trigonometry is shown for the ray optic interpretation of the incident electric field (E) of the charge terminal T₁ with a complex Brewster angle (θ_(i,B)) at the Hankel crossover distance (R_(x)). Recall that, for a lossy conducting medium, the Brewster angle is complex and specified by equation (38). Electrically, the geometric parameters are related by the electrical effective height (h_(eff)) of the charge terminal T₁ by equation (39). Since both the physical height (h_(p)) and the Hankel crossover distance (R_(x)) are real quantities, the angle of the desired guided surface wave tilt at the Hankel crossover distance (W_(Rx)) is equal to the phase (Φ) of the complex effective height (h_(eff)). With the charge terminal T₁ positioned at the physical height h_(p) and excited with a charge having the appropriate phase Φ, the resulting electric field is incident with the lossy conducting medium boundary interface at the Hankel crossover distance R_(x), and at the Brewster angle. Under these conditions, the guided surface waveguide mode can be excited without reflection or substantially negligible reflection.

However, Equation (39) means that the physical height of the guided surface waveguide probe 200 can be relatively small. While this will excite the guided surface waveguide mode, this can result in an unduly large bound charge with little free charge. To compensate, the charge terminal T₁ can be raised to an appropriate elevation to increase the amount of free charge. As one example rule of thumb, the charge terminal T₁ can be positioned at an elevation of about 4-5 times (or more) the effective diameter of the charge terminal T₁. FIG. 6 illustrates the effect of raising the charge terminal T₁ above the physical height (h_(p)) shown in FIG. 5A. The increased elevation causes the distance at which the wave tilt is incident with the lossy conductive medium to move beyond the Hankel crossover point 121 (FIG. 5A). To improve coupling in the guided surface waveguide mode, and thus provide for a greater launching efficiency of the guided surface wave, a lower compensation terminal T₂ can be used to adjust the total effective height (h_(TE)) of the charge terminal T₁ such that the wave tilt at the Hankel crossover distance is at the Brewster angle.

Referring to FIG. 12, shown is an example of a guided surface waveguide probe 200 c that includes an elevated charge terminal T₁ and a lower compensation terminal T₂ that are arranged along a vertical axis z that is normal to a plane presented by the lossy conducting medium 203. In this respect, the charge terminal T₁ is placed directly above the compensation terminal T₂ although it is possible that some other arrangement of two or more charge and/or compensation terminals T_(N) can be used. The guided surface waveguide probe 200 c is disposed above a lossy conducting medium 203 according to an embodiment of the present disclosure. The lossy conducting medium 203 makes up Region 1 with a second medium 206 that makes up Region 2 sharing a boundary interface with the lossy conducting medium 203.

The guided surface waveguide probe 200 c includes a feed network 209 that couples an excitation source 212 to the charge terminal T₁ and the compensation terminal T₂. According to various embodiments, charges Q₁ and Q₂ can be imposed on the respective charge and compensation terminals T₁ and T₂, depending on the voltages applied to terminals T₁ and T₂ at any given instant. I₁ is the conduction current feeding the charge Q₁ on the charge terminal T₁ via the terminal lead, and I₂ is the conduction current feeding the charge Q₂ on the compensation terminal T₂ via the terminal lead.

According to the embodiment of FIG. 12, the charge terminal T₁ is positioned over the lossy conducting medium 203 at a physical height H₁, and the compensation terminal T₂ is positioned directly below T₁ along the vertical axis z at a physical height H₂, where H₂ is less than H₁. The height h of the transmission structure may be calculated as h=H₁-H₂. The charge terminal T₁ has an isolated (or self) capacitance C₁, and the compensation terminal T₂ has an isolated (or self) capacitance C₂. A mutual capacitance C_(M) can also exist between the terminals T₁ and T₂ depending on the distance therebetween. During operation, charges Q₁ and Q₂ are imposed on the charge terminal T₁ and the compensation terminal T₂, respectively, depending on the voltages applied to the charge terminal T₁ and the compensation terminal T₂ at any given instant.

Referring next to FIG. 13, shown is a ray optics interpretation of the effects produced by the elevated charge Q₁ on charge terminal T₁ and compensation terminal T₂ of FIG. 12. With the charge terminal T₁ elevated to a height where the ray intersects with the lossy conductive medium at the Brewster angle at a distance greater than the Hankel crossover point 121 as illustrated by line 163, the compensation terminal T₂ can be used to adjust h_(TE) by compensating for the increased height. The effect of the compensation terminal T₂ is to reduce the electrical effective height of the guided surface waveguide probe (or effectively raise the lossy medium interface) such that the wave tilt at the Hankel crossover distance is at the Brewster angle as illustrated by line 166.

The total effective height can be written as the superposition of an upper effective height (h_(UE)) associated with the charge terminal T₁ and a lower effective height (h_(LE)) associated with the compensation terminal T₂ such that

h _(TE) =h _(UE) +h _(LE) =h _(p) e ^(j(βh) ^(p) ^(+Φ) ^(U) ⁾ +h _(d) e ^(j(βh) ^(d) ^(+Φ) ^(L) ⁾ =R _(x) ×W,  (85)

where Φ_(U) is the phase delay applied to the upper charge terminal T₁, Φ_(L) is the phase delay applied to the lower compensation terminal T₂, β=2π/λ_(p) is the propagation factor from Equation (35), h_(p) is the physical height of the charge terminal T₁ and h_(d) is the physical height of the compensation terminal T₂. If extra lead lengths are taken into consideration, they can be accounted for by adding the charge terminal lead length z to the physical height h_(p) of the charge terminal T₁ and the compensation terminal lead length y to the physical height h_(d) of the compensation terminal T₂ as shown in

h _(TE)=(h _(p) +z)e ^(j(β(h) ^(p) ^(+z)+Φ) ^(U) ⁾+(h _(d) +y)e ^(j(β(h) ^(d) ^(+y)+Φ) ^(L) ⁾ =R _(x) ×W.  (86)

The lower effective height can be used to adjust the total effective height (h_(TE)) to equal the complex effective height (h_(eff)) of FIG. 5A.

Equations (85) or (86) can be used to determine the physical height of the lower disk of the compensation terminal T₂ and the phase angles to feed the terminals in order to obtain the desired wave tilt at the Hankel crossover distance. For example, Equation (86) can be rewritten as the phase shift applied to the charge terminal T₁ as a function of the compensation terminal height (h_(d)) to give

$\begin{matrix} {{\Phi_{U}\left( h_{d} \right)} = {{- {\beta \left( {h_{p} + z} \right)}} - {j\mspace{11mu} {{\ln\left( \frac{{R_{x} \times W} - {\left( {h_{d} + y} \right)e^{j{({{\beta \; h_{d}} + {\beta \; y} + \Phi_{L}})}}}}{\left( {h_{p} + z} \right)} \right)}.}}}} & (87) \end{matrix}$

To determine the positioning of the compensation terminal T₂, the relationships discussed above can be utilized. First, the total effective height (h_(TE)) is the superposition of the complex effective height (h_(UE)) of the upper charge terminal T₁ and the complex effective height (h_(LE)) of the lower compensation terminal T₂ as expressed in Equation (86). Next, the tangent of the angle of incidence can be expressed geometrically as

$\begin{matrix} {{{\tan \mspace{11mu} \psi_{E}} = \frac{h_{TE}}{R_{x}}},} & (88) \end{matrix}$

which is equal to the definition of the wave tilt, W. Finally, given the desired Hankel crossover distance R_(x), the h_(TE) can be adjusted to make the wave tilt of the incident ray match the complex Brewster angle at the Hankel crossover point 121. This can be accomplished by adjusting h_(p), Φ_(U), and/or h_(d).

These concepts may be better understood when discussed in the context of an example of a guided surface waveguide probe. Referring to FIG. 14, shown is a graphical representation of an example of a guided surface waveguide probe 200 d including an upper charge terminal T₁ (e.g., a sphere at height h_(T)) and a lower compensation terminal T₂ (e.g., a disk at height h_(d)) that are positioned along a vertical axis z that is substantially normal to the plane presented by the lossy conducting medium 203. During operation, charges Q₁ and Q₂ are imposed on the charge and compensation terminals T₁ and T₂, respectively, depending on the voltages applied to the terminals T₁ and T₂ at any given instant.

An AC source 212 acts as the excitation source for the charge terminal T₁, which is coupled to the guided surface waveguide probe 200 d through a feed network 209 comprising a coil 215 such as, e.g., a helical coil. The AC source 212 can be connected across a lower portion of the coil 215 through a tap 227, as shown in FIG. 14, or can be inductively coupled to the coil 215 by way of a primary coil. The coil 215 can be coupled to a ground stake 218 at a first end and the charge terminal T₁ at a second end. In some implementations, the connection to the charge terminal T₁ can be adjusted using a tap 224 at the second end of the coil 215. The compensation terminal T₂ is positioned above and substantially parallel with the lossy conducting medium 203 (e.g., the ground or Earth), and energized through a tap 233 coupled to the coil 215. An ammeter 236 located between the coil 215 and ground stake 218 can be used to provide an indication of the magnitude of the current flow (I₀) at the base of the guided surface waveguide probe. Alternatively, a current clamp may be used around the conductor coupled to the ground stake 218 to obtain an indication of the magnitude of the current flow (I₀).

In the example of FIG. 14, the coil 215 is coupled to a ground stake 218 at a first end and the charge terminal T₁ at a second end via a vertical feed line conductor 221. In some implementations, the connection to the charge terminal T₁ can be adjusted using a tap 224 at the second end of the coil 215 as shown in FIG. 14. The coil 215 can be energized at an operating frequency by the AC source 212 through a tap 227 at a lower portion of the coil 215. In other implementations, the AC source 212 can be inductively coupled to the coil 215 through a primary coil. The compensation terminal T₂ is energized through a tap 233 coupled to the coil 215. An ammeter 236 located between the coil 215 and ground stake 218 can be used to provide an indication of the magnitude of the current flow at the base of the guided surface waveguide probe 200 d. Alternatively, a current clamp may be used around the conductor coupled to the ground stake 218 to obtain an indication of the magnitude of the current flow. The compensation terminal T₂ is positioned above and substantially parallel with the lossy conducting medium 203 (e.g., the ground).

In the example of FIG. 14, the connection to the charge terminal T₁ located on the coil 215 above the connection point of tap 233 for the compensation terminal T₂. Such an adjustment allows an increased voltage (and thus a higher charge Q₁) to be applied to the upper charge terminal T₁. In other embodiments, the connection points for the charge terminal T₁ and the compensation terminal T₂ can be reversed. It is possible to adjust the total effective height (h_(TE)) of the guided surface waveguide probe 200 d to excite an electric field having a guided surface wave tilt at the Hankel crossover distance R_(x). The Hankel crossover distance can also be found by equating the magnitudes of equations (20b) and (21) for −jγρ, and solving for R_(x) as illustrated by FIG. 4. The index of refraction (n), the complex Brewster angle (θ_(i,B) and ψ_(i,B)), the wave tilt (|W|e^(jΨ)) and the complex effective height (h_(eff)=h_(p)e^(jΦ)) can be determined as described with respect to Equations (41)-(44) above.

With the selected charge terminal T₁ configuration, a spherical diameter (or the effective spherical diameter) can be determined. For example, if the charge terminal T₁ is not configured as a sphere, then the terminal configuration may be modeled as a spherical capacitance having an effective spherical diameter. The size of the charge terminal T₁ can be chosen to provide a sufficiently large surface for the charge Q₁ imposed on the terminals. In general, it is desirable to make the charge terminal T₁ as large as practical. The size of the charge terminal T₁ should be large enough to avoid ionization of the surrounding air, which can result in electrical discharge or sparking around the charge terminal. To reduce the amount of bound charge on the charge terminal T₁, the desired elevation to provide free charge on the charge terminal T₁ for launching a guided surface wave should be at least 4-5 times the effective spherical diameter above the lossy conductive medium (e.g., the Earth). The compensation terminal T₂ can be used to adjust the total effective height (h_(TE)) of the guided surface waveguide probe 200 d to excite an electric field having a guided surface wave tilt at R_(x). The compensation terminal T₂ can be positioned below the charge terminal T₁ at h_(d)=h_(T)−h_(p), where h_(T) is the total physical height of the charge terminal T₁. With the position of the compensation terminal T₂ fixed and the phase delay Φ_(U) applied to the upper charge terminal T₁, the phase delay Φ_(L) applied to the lower compensation terminal T₂ can be determined using the relationships of Equation (86), such that:

$\begin{matrix} {{\Phi_{U}\left( h_{d} \right)} = {{- {\beta \left( {h_{d} + y} \right)}} - {j\mspace{11mu} {{\ln\left( \frac{{R_{x} \times W} - {\left( {h_{p} + z} \right)e^{j{({{\beta \; h_{p}} + {\beta \; z} + \Phi_{L}})}}}}{\left( {h_{d} + y} \right)} \right)}.}}}} & (89) \end{matrix}$

In alternative embodiments, the compensation terminal T₂ can be positioned at a height h_(d) where Im{Φ_(L)}=0. This is graphically illustrated in FIG. 15A, which shows plots 172 and 175 of the imaginary and real parts of Φ_(U), respectively. The compensation terminal T₂ is positioned at a height h_(d) where Im{Φ_(U)}=0, as graphically illustrated in plot 172. At this fixed height, the coil phase Φ_(U) can be determined from Re{Φ_(U)}, as graphically illustrated in plot 175.

With the AC source 212 coupled to the coil 215 (e.g., at the 50Ω point to maximize coupling), the position of tap 233 may be adjusted for parallel resonance of the compensation terminal T₂ with at least a portion of the coil at the frequency of operation. FIG. 15B shows a schematic diagram of the general electrical hookup of FIG. 14 in which V₁ is the voltage applied to the lower portion of the coil 215 from the AC source 212 through tap 227, V₂ is the voltage at tap 224 that is supplied to the upper charge terminal T₁, and V₃ is the voltage applied to the lower compensation terminal T₂ through tap 233. The resistances R_(p) and R_(d) represent the ground return resistances of the charge terminal T₁ and compensation terminal T₂, respectively. The charge and compensation terminals T₁ and T₂ may be configured as spheres, cylinders, toroids, rings, hoods, or any other combination of capacitive structures. The size of the charge and compensation terminals T₁ and T₂ can be chosen to provide a sufficiently large surface for the charges Q₁ and Q₂ imposed on the terminals. In general, it is desirable to make the charge terminal T₁ as large as practical. The size of the charge terminal T₁ should be large enough to avoid ionization of the surrounding air, which can result in electrical discharge or sparking around the charge terminal. The self-capacitance C_(p) and C_(d) of the charge and compensation terminals T₁ and T₂ respectively, can be determined using, for example, equation (24).

As can be seen in FIG. 15B, a resonant circuit is formed by at least a portion of the inductance of the coil 215, the self-capacitance C_(d) of the compensation terminal T₂, and the ground return resistance R_(d) associated with the compensation terminal T₂. The parallel resonance can be established by adjusting the voltage V₃ applied to the compensation terminal T₂ (e.g., by adjusting a tap 233 position on the coil 215) or by adjusting the height and/or size of the compensation terminal T₂ to adjust C_(d). The position of the coil tap 233 can be adjusted for parallel resonance, which will result in the ground current through the ground stake 218 and through the ammeter 236 reaching a maximum point. After parallel resonance of the compensation terminal T₂ has been established, the position of the tap 227 for the AC source 212 can be adjusted to the 50Ω point on the coil 215.

Voltage V₂ from the coil 215 can be applied to the charge terminal T₁, and the position of tap 224 can be adjusted such that the phase (Φ) of the total effective height (h_(TE)) approximately equals the angle of the guided surface wave tilt (W_(Rx)) at the Hankel crossover distance (R_(x)). The position of the coil tap 224 can be adjusted until this operating point is reached, which results in the ground current through the ammeter 236 increasing to a maximum. At this point, the resultant fields excited by the guided surface waveguide probe 200 d are substantially mode-matched to a guided surface waveguide mode on the surface of the lossy conducting medium 203, resulting in the launching of a guided surface wave along the surface of the lossy conducting medium 203. This can be verified by measuring field strength along a radial extending from the guided surface waveguide probe 200.

Resonance of the circuit including the compensation terminal T₂ may change with the attachment of the charge terminal T₁ and/or with adjustment of the voltage applied to the charge terminal T₁ through tap 224. While adjusting the compensation terminal circuit for resonance aids the subsequent adjustment of the charge terminal connection, it is not necessary to establish the guided surface wave tilt (W_(Rx)) at the Hankel crossover distance (R_(x)). The system may be further adjusted to improve coupling by iteratively adjusting the position of the tap 227 for the AC source 212 to be at the 50Ω point on the coil 215 and adjusting the position of tap 233 to maximize the ground current through the ammeter 236. Resonance of the circuit including the compensation terminal T₂ may drift as the positions of taps 227 and 233 are adjusted, or when other components are attached to the coil 215.

In other implementations, the voltage V₂ from the coil 215 can be applied to the charge terminal T₁, and the position of tap 233 can be adjusted such that the phase (Φ) of the total effective height (h_(TE)) approximately equals the angle (Ψ) of the guided surface wave tilt at R_(x). The position of the coil tap 224 can be adjusted until the operating point is reached, resulting in the ground current through the ammeter 236 substantially reaching a maximum. The resultant fields are substantially mode-matched to a guided surface waveguide mode on the surface of the lossy conducting medium 203, and a guided surface wave is launched along the surface of the lossy conducting medium 203. This can be verified by measuring field strength along a radial extending from the guided surface waveguide probe 200. The system may be further adjusted to improve coupling by iteratively adjusting the position of the tap 227 for the AC source 212 to be at the 50Ω point on the coil 215 and adjusting the position of tap 224 and/or 233 to maximize the ground current through the ammeter 236.

Referring back to FIG. 12, operation of a guided surface waveguide probe 200 may be controlled to adjust for variations in operational conditions associated with the guided surface waveguide probe 200. For example, a probe control system 230 can be used to control the feed network 209 and/or positioning of the charge terminal T₁ and/or compensation terminal T₂ to control the operation of the guided surface waveguide probe 200. Operational conditions can include, but are not limited to, variations in the characteristics of the lossy conducting medium 203 (e.g., conductivity σ and relative permittivity ε_(r)), variations in field strength and/or variations in loading of the guided surface waveguide probe 200. As can be seen from Equations (41)-(44), the index of refraction (n), the complex Brewster angle (θ_(i,B) and ψ_(i,B)), the wave tilt (|W|e^(jΨ)) and the complex effective height (h_(eff)=h_(p)e^(jΦ)) can be affected by changes in soil conductivity and permittivity resulting from, e.g., weather conditions.

Equipment such as, e.g., conductivity measurement probes, permittivity sensors, ground parameter meters, field meters, current monitors and/or load receivers can be used to monitor for changes in the operational conditions and provide information about current operational conditions to the probe control system 230. The probe control system 230 can then make one or more adjustments to the guided surface waveguide probe 200 to maintain specified operational conditions for the guided surface waveguide probe 200. For instance, as the moisture and temperature vary, the conductivity of the soil will also vary. Conductivity measurement probes and/or permittivity sensors may be located at multiple locations around the guided surface waveguide probe 200. Generally, it would be desirable to monitor the conductivity and/or permittivity at or about the Hankel crossover distance R_(x) for the operational frequency. Conductivity measurement probes and/or permittivity sensors may be located at multiple locations (e.g., in each quadrant) around the guided surface waveguide probe 200.

With reference then to FIG. 16, shown is an example of a guided surface waveguide probe 200 e that includes a charge terminal T₁ and a charge terminal T₂ that are arranged along a vertical axis z. The guided surface waveguide probe 200 e is disposed above a lossy conducting medium 203, which makes up Region 1. In addition, a second medium 206 shares a boundary interface with the lossy conducting medium 203 and makes up Region 2. The charge terminals T₁ and T₂ are positioned over the lossy conducting medium 203. The charge terminal T₁ is positioned at height H₁, and the charge terminal T₂ is positioned directly below T₁ along the vertical axis z at height H₂, where H₂ is less than H₁. The height h of the transmission structure presented by the guided surface waveguide probe 200 e is h=H₁-H₂. The guided surface waveguide probe 200 e includes a feed network 209 that couples an excitation source 212 to the charge terminals T₁ and T₂.

The charge terminals T₁ and/or T₂ include a conductive mass that can hold an electrical charge, which may be sized to hold as much charge as practically possible. The charge terminal T₁ has a self-capacitance C₁, and the charge terminal T₂ has a self-capacitance C₂, which can be determined using, for example, equation (24). By virtue of the placement of the charge terminal T₁ directly above the charge terminal T₂, a mutual capacitance C_(M) is created between the charge terminals T₁ and T₂. Note that the charge terminals T₁ and T₂ need not be identical, but each can have a separate size and shape, and can include different conducting materials. Ultimately, the field strength of a guided surface wave launched by a guided surface waveguide probe 200 e is directly proportional to the quantity of charge on the terminal T₁. The charge Q₁ is, in turn, proportional to the self-capacitance C₁ associated with the charge terminal T₁ since Q₁=C₁V, where V is the voltage imposed on the charge terminal T₁.

When properly adjusted to operate at a predefined operating frequency, the guided surface waveguide probe 200 e generates a guided surface wave along the surface of the lossy conducting medium 203. The excitation source 212 can generate electrical energy at the predefined frequency that is applied to the guided surface waveguide probe 200 e to excite the structure. When the electromagnetic fields generated by the guided surface waveguide probe 200 e are substantially mode-matched with the lossy conducting medium 203, the electromagnetic fields substantially synthesize a wave front incident at a complex Brewster angle that results in little or no reflection. Thus, the surface waveguide probe 200 e does not produce a radiated wave, but launches a guided surface traveling wave along the surface of a lossy conducting medium 203. The energy from the excitation source 212 can be transmitted as Zenneck surface currents to one or more receivers that are located within an effective transmission range of the guided surface waveguide probe 200 e.

One can determine asymptotes of the radial Zenneck surface current J_(ρ) (ρ) on the surface of the lossy conducting medium 203 to be J₁(ρ) close-in and J₂(ρ) far-out, where

$\begin{matrix} {{{{Close}\text{-}{in}\mspace{11mu} \left( {\rho < {\lambda/8}} \right)\text{:}\mspace{14mu} {\left. {J_{\rho}(\rho)} \right.\sim J_{1}}} = {\frac{I_{1} + I_{2}}{2{\pi\rho}} + \frac{{E_{\rho}^{QS}\left( Q_{1} \right)} + {E_{\rho}^{QS}\left( Q_{2} \right)}}{Z_{\rho}}}},{and}} & (90) \\ {\mspace{79mu} {{{Far}\text{-}{out}\mspace{14mu} \left( {\rho\operatorname{>>}{\lambda/8}} \right)\text{:}\mspace{14mu} {\left. {J_{\rho}(\rho)} \right.\sim J_{2}}} = {\frac{j\; {\gamma\omega}\; Q_{1}}{4} \times \sqrt{\frac{2\gamma}{\pi}} \times {\frac{e^{{- {({\alpha + {j\; \beta}})}}\rho}}{\sqrt{\rho}}.}}}} & (91) \end{matrix}$

where I₁ is the conduction current feeding the charge Q₁ on the first charge terminal T₁, and I₂ is the conduction current feeding the charge Q₂ on the second charge terminal T₂. The charge Q₁ on the upper charge terminal T₁ is determined by Q₁=C₁V₁, where C₁ is the isolated capacitance of the charge terminal T₁. Note that there is a third component to J₁ set forth above given by (E_(ρ) ^(Q) ¹ )/Z_(ρ), which follows from the Leontovich boundary condition and is the radial current contribution in the lossy conducting medium 203 pumped by the quasi-static field of the elevated oscillating charge on the first charge terminal Q₁. The quantity Z_(ρ)=jωμ_(o)/γ_(e) is the radial impedance of the lossy conducting medium, where γ_(e)=(jωμ₁σ₁−ω²μ₁ε₁)^(1/2).

The asymptotes representing the radial current close-in and far-out as set forth by equations (90) and (91) are complex quantities. According to various embodiments, a physical surface current J(ρ), is synthesized to match as close as possible the current asymptotes in magnitude and phase. That is to say close-in, |J(ρ)| is to be tangent to |J₁|, and far-out |J(ρ)| is to be tangent to |J₂|. Also, according to the various embodiments, the phase of J(ρ) should transition from the phase of J₁ close-in to the phase of J₂ far-out.

In order to match the guided surface wave mode at the site of transmission to launch a guided surface wave, the phase of the surface current |J₂| far-out should differ from the phase of the surface current |J₁| close-in by the propagation phase corresponding to e^(−jβ(ρ) ² ^(−ρ) ¹ ⁾ plus a constant of approximately 45 degrees or 225 degrees. This is because there are two roots for √{square root over (γ)}, one near π/4 and one near 5π/4. The properly adjusted synthetic radial surface current is

$\begin{matrix} {{J_{\rho}\left( {\rho,\varphi,0} \right)} = {\frac{I_{o}\gamma}{4}{{H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}.}}} & (92) \end{matrix}$

Note that this is consistent with equation (17). By Maxwell's equations, such a J(φ surface current automatically creates fields that conform to

$\begin{matrix} {{H_{\varphi} = {\frac{{- \gamma}\; I_{o}}{4}e^{{- u_{2}}z}\mspace{11mu} {H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},} & (93) \\ {{E_{\rho} = {\frac{{- \gamma}\; I_{o}}{4}\left( \frac{u_{2}}{j\; {\omega ɛ}_{o}} \right)e^{{- u_{2}}z}\mspace{11mu} {H_{1}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}}},{and}} & (94) \\ {E_{z} = {\frac{{- \gamma}\; I_{o}}{4}\left( \frac{- \gamma}{{\omega ɛ}_{o}} \right)e^{{- u_{2}}z}\mspace{11mu} {{H_{0}^{(2)}\left( {{- j}\; {\gamma\rho}} \right)}.}}} & (95) \end{matrix}$

Thus, the difference in phase between the surface current |J₂| far-out and the surface current |J₁| close-in for the guided surface wave mode that is to be matched is due to the characteristics of the Hankel functions in equations (93)-(95), which are consistent with equations (1)-(3). It is of significance to recognize that the fields expressed by equations (1)-(6) and (17) and equations (92)-(95) have the nature of a transmission line mode bound to a lossy interface, not radiation fields that are associated with groundwave propagation.

In order to obtain the appropriate voltage magnitudes and phases for a given design of a guided surface waveguide probe 200 e at a given location, an iterative approach may be used. Specifically, analysis may be performed of a given excitation and configuration of a guided surface waveguide probe 200 e taking into account the feed currents to the terminals T₁ and T₂, the charges on the charge terminals T₁ and T₂, and their images in the lossy conducting medium 203 in order to determine the radial surface current density generated. This process may be performed iteratively until an optimal configuration and excitation for a given guided surface waveguide probe 200 e is determined based on desired parameters. To aid in determining whether a given guided surface waveguide probe 200 e is operating at an optimal level, a guided field strength curve 103 (FIG. 1) may be generated using equations (1)-(12) based on values for the conductivity of Region 1 (σ₁) and the permittivity of Region 1 (ε₁) at the location of the guided surface waveguide probe 200 e. Such a guided field strength curve 103 can provide a benchmark for operation such that measured field strengths can be compared with the magnitudes indicated by the guided field strength curve 103 to determine if optimal transmission has been achieved.

In order to arrive at an optimized condition, various parameters associated with the guided surface waveguide probe 200 e may be adjusted. One parameter that may be varied to adjust the guided surface waveguide probe 200 e is the height of one or both of the charge terminals T₁ and/or T₂ relative to the surface of the lossy conducting medium 203. In addition, the distance or spacing between the charge terminals T₁ and T₂ may also be adjusted. In doing so, one may minimize or otherwise alter the mutual capacitance C_(M) or any bound capacitances between the charge terminals T₁ and T₂ and the lossy conducting medium 203 as can be appreciated. The size of the respective charge terminals T₁ and/or T₂ can also be adjusted. By changing the size of the charge terminals T₁ and/or T₂, one will alter the respective self-capacitances C₁ and/or C₂, and the mutual capacitance C_(M) as can be appreciated.

Still further, another parameter that can be adjusted is the feed network 209 associated with the guided surface waveguide probe 200 e. This may be accomplished by adjusting the size of the inductive and/or capacitive reactances that make up the feed network 209. For example, where such inductive reactances comprise coils, the number of turns on such coils may be adjusted. Ultimately, the adjustments to the feed network 209 can be made to alter the electrical length of the feed network 209, thereby affecting the voltage magnitudes and phases on the charge terminals T₁ and T₂.

Note that the iterations of transmission performed by making the various adjustments may be implemented by using computer models or by adjusting physical structures as can be appreciated. By making the above adjustments, one can create corresponding “close-in” surface current J₁ and “far-out” surface current J₂ that approximate the same currents J(ρ) of the guided surface wave mode specified in Equations (90) and (91) set forth above. In doing so, the resulting electromagnetic fields would be substantially or approximately mode-matched to a guided surface wave mode on the surface of the lossy conducting medium 203.

While not shown in the example of FIG. 16, operation of the guided surface waveguide probe 200 e may be controlled to adjust for variations in operational conditions associated with the guided surface waveguide probe 200. For example, a probe control system 230 shown in FIG. 12 can be used to control the feed network 209 and/or positioning and/or size of the charge terminals T₁ and/or T₂ to control the operation of the guided surface waveguide probe 200 e. Operational conditions can include, but are not limited to, variations in the characteristics of the lossy conducting medium 203 (e.g., conductivity a and relative permittivity ε_(r)), variations in field strength and/or variations in loading of the guided surface waveguide probe 200 e.

Referring now to FIG. 17, shown is an example of the guided surface waveguide probe 200 e of FIG. 16, denoted herein as guided surface waveguide probe 200 f. The guided surface waveguide probe 200 f includes the charge terminals T₁ and T₂ that are positioned along a vertical axis z that is substantially normal to the plane presented by the lossy conducting medium 203 (e.g., the Earth). The second medium 206 is above the lossy conducting medium 203. The charge terminal T₁ has a self-capacitance C₁, and the charge terminal T₂ has a self-capacitance C₂. During operation, charges Q₁ and Q₂ are imposed on the charge terminals T₁ and T₂, respectively, depending on the voltages applied to the charge terminals T₁ and T₂ at any given instant. A mutual capacitance C_(M) may exist between the charge terminals T₁ and T₂ depending on the distance there between. In addition, bound capacitances may exist between the respective charge terminals T₁ and T₂ and the lossy conducting medium 203 depending on the heights of the respective charge terminals T₁ and T₂ with respect to the lossy conducting medium 203.

The guided surface waveguide probe 200 f includes a feed network 209 that comprises an inductive impedance comprising a coil L_(1a) having a pair of leads that are coupled to respective ones of the charge terminals T₁ and T₂. In one embodiment, the coil L_(1a) is specified to have an electrical length that is one-half (½) of the wavelength at the operating frequency of the guided surface waveguide probe 200 f.

While the electrical length of the coil L_(1a) is specified as approximately one-half (½) the wavelength at the operating frequency, it is understood that the coil L_(1a) may be specified with an electrical length at other values. According to one embodiment, the fact that the coil L_(1a) has an electrical length of approximately one-half the wavelength at the operating frequency provides for an advantage in that a maximum voltage differential is created on the charge terminals T₁ and T₂. Nonetheless, the length or diameter of the coil L_(1a) may be increased or decreased when adjusting the guided surface waveguide probe 200 f to obtain optimal excitation of a guided surface wave mode. Adjustment of the coil length may be provided by taps located at one or both ends of the coil. In other embodiments, it may be the case that the inductive impedance is specified to have an electrical length that is significantly less than or greater than ½ the wavelength at the operating frequency of the guided surface waveguide probe 200 f.

The excitation source 212 can be coupled to the feed network 209 by way of magnetic coupling. Specifically, the excitation source 212 is coupled to a coil L_(P) that is inductively coupled to the coil L_(1a). This may be done by link coupling, a tapped coil, a variable reactance, or other coupling approach as can be appreciated. To this end, the coil L_(P) acts as a primary, and the coil L_(1a) acts as a secondary as can be appreciated.

In order to adjust the guided surface waveguide probe 200 f for the transmission of a desired guided surface wave, the heights of the respective charge terminals T₁ and T₂ may be altered with respect to the lossy conducting medium 203 and with respect to each other. Also, the sizes of the charge terminals T₁ and T₂ may be altered. In addition, the size of the coil L_(1a) may be altered by adding or eliminating turns or by changing some other dimension of the coil L_(1a). The coil L_(1a) can also include one or more taps for adjusting the electrical length as shown in FIG. 17. The position of a tap connected to either charge terminal T₁ or T₂ can also be adjusted.

Referring next to FIGS. 18A, 18B, 18C and 19, shown are examples of generalized receive circuits for using the surface-guided waves in wireless power delivery systems. FIGS. 18A and 18B-18C include a linear probe 303 and a tuned resonator 306, respectively. FIG. 19 is a magnetic coil 309 according to various embodiments of the present disclosure. According to various embodiments, each one of the linear probe 303, the tuned resonator 306, and the magnetic coil 309 may be employed to receive power transmitted in the form of a guided surface wave on the surface of a lossy conducting medium 203 according to various embodiments. As mentioned above, in one embodiment the lossy conducting medium 203 comprises a terrestrial medium (or Earth).

With specific reference to FIG. 18A, the open-circuit terminal voltage at the output terminals 312 of the linear probe 303 depends upon the effective height of the linear probe 303. To this end, the terminal point voltage may be calculated as

V _(T)=∫₀ ^(h) ^(e) E _(inc) ·dl,  (96)

where E_(inc) is the strength of the incident electric field induced on the linear probe 303 in Volts per meter, dl is an element of integration along the direction of the linear probe 303, and h_(e) is the effective height of the linear probe 303. An electrical load 315 is coupled to the output terminals 312 through an impedance matching network 318.

When the linear probe 303 is subjected to a guided surface wave as described above, a voltage is developed across the output terminals 312 that may be applied to the electrical load 315 through a conjugate impedance matching network 318 as the case may be. In order to facilitate the flow of power to the electrical load 315, the electrical load 315 should be substantially impedance matched to the linear probe 303 as will be described below.

Referring to FIG. 18B, a ground current excited coil 306 a possessing a phase shift equal to the wave tilt of the guided surface wave includes a charge terminal T_(R) that is elevated (or suspended) above the lossy conducting medium 203. The charge terminal T_(R) has a self-capacitance C_(R). In addition, there may also be a bound capacitance (not shown) between the charge terminal T_(R) and the lossy conducting medium 203 depending on the height of the charge terminal T_(R) above the lossy conducting medium 203. The bound capacitance should preferably be minimized as much as is practicable, although this may not be entirely necessary in every instance.

The tuned resonator 306 a also includes a receiver network comprising a coil L_(R) having a phase shift Φ. One end of the coil L_(R) is coupled to the charge terminal T_(R), and the other end of the coil L_(R) is coupled to the lossy conducting medium 203. The receiver network can include a vertical supply line conductor that couples the coil L_(R) to the charge terminal T_(R). To this end, the coil L_(R) (which may also be referred to as tuned resonator L_(R)-C_(R)) comprises a series-adjusted resonator as the charge terminal C_(R) and the coil L_(R) are situated in series. The phase delay of the coil L_(R) can be adjusted by changing the size and/or height of the charge terminal T_(R), and/or adjusting the size of the coil L_(R) so that the phase Φ of the structure is made substantially equal to the angle of the wave tilt Ψ. The phase delay of the vertical supply line can also be adjusted by, e.g., changing length of the conductor.

For example, the reactance presented by the self-capacitance C_(R) is calculated as 1/jωC_(R). Note that the total capacitance of the structure 306 a may also include capacitance between the charge terminal T_(R) and the lossy conducting medium 203, where the total capacitance of the structure 306 a may be calculated from both the self-capacitance C_(R) and any bound capacitance as can be appreciated. According to one embodiment, the charge terminal T_(R) may be raised to a height so as to substantially reduce or eliminate any bound capacitance. The existence of a bound capacitance may be determined from capacitance measurements between the charge terminal T_(R) and the lossy conducting medium 203 as previously discussed.

The inductive reactance presented by a discrete-element coil L_(R) may be calculated as jωL, where L is the lumped-element inductance of the coil L_(R). If the coil L_(R) is a distributed element, its equivalent terminal-point inductive reactance may be determined by conventional approaches. To tune the structure 306 a, one would make adjustments so that the phase delay is equal to the wave tilt for the purpose of mode-matching to the surface waveguide at the frequency of operation. Under this condition, the receiving structure may be considered to be “mode-matched” with the surface waveguide. A transformer link around the structure and/or an impedance matching network 324 may be inserted between the probe and the electrical load 327 in order to couple power to the load. Inserting the impedance matching network 324 between the probe terminals 321 and the electrical load 327 can effect a conjugate-match condition for maximum power transfer to the electrical load 327.

When placed in the presence of surface currents at the operating frequencies power will be delivered from the surface guided wave to the electrical load 327. To this end, an electrical load 327 may be coupled to the structure 306 a by way of magnetic coupling, capacitive coupling, or conductive (direct tap) coupling. The elements of the coupling network may be lumped components or distributed elements as can be appreciated.

In the embodiment shown in FIG. 18B, magnetic coupling is employed where a coil L_(S) is positioned as a secondary relative to the coil L_(R) that acts as a transformer primary. The coil L_(S) may be link-coupled to the coil L_(R) by geometrically winding it around the same core structure and adjusting the coupled magnetic flux as can be appreciated. In addition, while the receiving structure 306 a comprises a series-tuned resonator, a parallel-tuned resonator or even a distributed-element resonator of the appropriate phase delay may also be used.

While a receiving structure immersed in an electromagnetic field may couple energy from the field, it can be appreciated that polarization-matched structures work best by maximizing the coupling, and conventional rules for probe-coupling to waveguide modes should be observed. For example, a TE₂₀ (transverse electric mode) waveguide probe may be optimal for extracting energy from a conventional waveguide excited in the TE₂₀ mode. Similarly, in these cases, a mode-matched and phase-matched receiving structure can be optimized for coupling power from a surface-guided wave. The guided surface wave excited by a guided surface waveguide probe 200 on the surface of the lossy conducting medium 203 can be considered a waveguide mode of an open waveguide. Excluding waveguide losses, the source energy can be completely recovered. Useful receiving structures may be E-field coupled, H-field coupled, or surface-current excited.

The receiving structure can be adjusted to increase or maximize coupling with the guided surface wave based upon the local characteristics of the lossy conducting medium 203 in the vicinity of the receiving structure. To accomplish this, the phase delay (Φ) of the receiving structure can be adjusted to match the angle (Ψ) of the wave tilt of the surface traveling wave at the receiving structure. If configured appropriately, the receiving structure may then be tuned for resonance with respect to the perfectly conducting image ground plane at complex depth z=−d/2.

For example, consider a receiving structure comprising the tuned resonator 306 a of FIG. 18B, including a coil L_(R) and a vertical supply line connected between the coil L_(R) and a charge terminal T_(R). With the charge terminal T_(R) positioned at a defined height above the lossy conducting medium 203, the total phase shift Φ of the coil L_(R) and vertical supply line can be matched with the angle (Ψ) of the wave tilt at the location of the tuned resonator 306 a. From Equation (22), it can be seen that the wave tilt asymptotically passes to

$\begin{matrix} {{W = {{{W}e^{j\; \Psi}} = {\frac{E_{\rho}}{E_{z}}\underset{\rho\rightarrow\infty}{\rightarrow}\frac{1}{\sqrt{ɛ_{r} - {j\; \frac{\sigma_{1}}{{\omega ɛ}_{o}}}}}}}},} & (97) \end{matrix}$

where ε_(r) comprises the relative permittivity and σ₁ is the conductivity of the lossy conducting medium 203 at the location of the receiving structure, ε_(o) is the permittivity of free space, and ω=2πf, where f is the frequency of excitation. Thus, the wave tilt angle (Ψ) can be determined from Equation (97).

The total phase shift (Φ=θ_(c)+θ_(y)) of the tuned resonator 306 a includes both the phase delay (θ_(c)) through the coil L_(R) and the phase delay of the vertical supply line (θ_(y)). The spatial phase delay along the conductor length l_(w) of the vertical supply line can be given by θ_(y)=β_(w)l_(w), where β_(w) is the propagation phase constant for the vertical supply line conductor. The phase delay due to the coil (or helical delay line) is θ_(c)=β_(p)l_(C), with a physical length of l_(C) and a propagation factor of

$\begin{matrix} {{\beta_{p} = {\frac{2\pi}{\lambda_{p}} = \frac{2\pi}{V_{f}\lambda_{0}}}},} & (98) \end{matrix}$

where V_(f) is the velocity factor on the structure, λ₀ is the wavelength at the supplied frequency, and λ_(p) is the propagation wavelength resulting from the velocity factor V_(f). One or both of the phase delays (θ_(c)+θ_(y)) can be adjusted to match the phase shift Φ to the angle (Ψ) of the wave tilt. For example, a tap position may be adjusted on the coil L_(R) of FIG. 18B to adjust the coil phase delay (θ_(c)) to match the total phase shift to the wave tilt angle (Φ=Ψ). For example, a portion of the coil can be bypassed by the tap connection as illustrated in FIG. 18B. The vertical supply line conductor can also be connected to the coil L_(R) via a tap, whose position on the coil may be adjusted to match the total phase shift to the angle of the wave tilt.

Once the phase delay (Φ) of the tuned resonator 306 a has been adjusted, the impedance of the charge terminal T_(R) can then be adjusted to tune to resonance with respect to the perfectly conducting image ground plane at complex depth z=−d/2. This can be accomplished by adjusting the capacitance of the charge terminal T₁ without changing the traveling wave phase delays of the coil L_(R) and vertical supply line. The adjustments are similar to those described with respect to FIGS. 9A and 9B.

The impedance seen “looking down” into the lossy conducting medium 203 to the complex image plane is given by:

Z _(in) =R _(in) +jX _(in) =Z _(o) tan h(jβ _(o)(d/2)),  (99)

where β_(o)=ω√{square root over (μ_(o)ε_(o))}. For vertically polarized sources over the Earth, the depth of the complex image plane can be given by:

d/2≈1/√{square root over (jωμ ₁σ₁−ω²μ₁ε₁)},  (100)

where μ₁ is the permeability of the lossy conducting medium 203 and ε₁=ε_(r)ε_(o).

At the base of the tuned resonator 306 a, the impedance seen “looking up” into the receiving structure is Z_(↑)=Z_(base) as illustrated in FIG. 9A. With a terminal impedance of:

$\begin{matrix} {{Z_{R} = \frac{1}{j\; \omega \; C_{R}}},} & (101) \end{matrix}$

where C_(R) is the self-capacitance of the charge terminal T_(R), the impedance seen “looking up” into the vertical supply line conductor of the tuned resonator 306 a is given by:

$\begin{matrix} {{Z_{2} = {{Z_{W}\frac{Z_{R} + {Z_{w}{\tanh \left( {j\; \beta_{w}h_{w}} \right)}}}{Z_{w} + {Z_{R}{\tanh \left( {j\; \beta_{w}h_{w}} \right)}}}} = {Z_{W}\frac{Z_{R} + {Z_{w}{\tanh \left( {j\; \theta_{y}} \right)}}}{Z_{w} + {Z_{R}{\tanh \left( {j\; \theta_{y}} \right)}}}}}},} & (102) \end{matrix}$

and the impedance seen “looking up” into the coil L_(R) of the tuned resonator 306 a is given by:

$\begin{matrix} {Z_{base} = {{R_{base} + {jX}_{base}} = {{Z_{R}\frac{Z_{2} + {Z_{R}{\tanh \left( {j\; \beta_{p}H} \right)}}}{Z_{R} + {Z_{2}{\tanh \left( {j\; \beta_{p}H} \right)}}}} = {Z_{c}{\frac{Z_{2} + {Z_{R}{\tanh \left( {j\; \theta_{c}} \right)}}}{Z_{R} + {Z_{2}{\tanh \left( {j\; \theta_{c}} \right)}}}.}}}}} & (103) \end{matrix}$

By matching the reactive component (X_(in)) seen “looking down” into the lossy conducting medium 203 with the reactive component (X_(base)) seen “looking up” into the tuned resonator 306 a, the coupling into the guided surface waveguide mode may be maximized.

Referring next to FIG. 180, shown is an example of a tuned resonator 306 b that does not include a charge terminal T_(R) at the top of the receiving structure. In this embodiment, the tuned resonator 306 b does not include a vertical supply line coupled between the coil L_(R) and the charge terminal T_(R). Thus, the total phase shift (Φ) of the tuned resonator 306 b includes only the phase delay (θ_(c)) through the coil L_(R). As with the tuned resonator 306 a of FIG. 18B, the coil phase delay θ_(c) can be adjusted to match the angle (Ψ) of the wave tilt determined from Equation (97), which results in Φ=Ψ. While power extraction is possible with the receiving structure coupled into the surface waveguide mode, it is difficult to adjust the receiving structure to maximize coupling with the guided surface wave without the variable reactive load provided by the charge terminal T_(R).

Referring to FIG. 18D, shown is a flow chart 180 illustrating an example of adjusting a receiving structure to substantially mode-match to a guided surface waveguide mode on the surface of the lossy conducting medium 203. Beginning with 181, if the receiving structure includes a charge terminal T_(R) (e.g., of the tuned resonator 306 a of FIG. 18B), then the charge terminal T_(R) is positioned at a defined height above a lossy conducting medium 203 at 184. As the surface guided wave has been established by a guided surface waveguide probe 200, the physical height (h_(p)) of the charge terminal T_(R) may be below that of the effective height. The physical height may be selected to reduce or minimize the bound charge on the charge terminal T_(R) (e.g., four times the spherical diameter of the charge terminal). If the receiving structure does not include a charge terminal T_(R) (e.g., of the tuned resonator 306 b of FIG. 18C), then the flow proceeds to 187.

At 187, the electrical phase delay Φ of the receiving structure is matched to the complex wave tilt angle Ψ defined by the local characteristics of the lossy conducting medium 203. The phase delay (θ_(c)) of the helical coil and/or the phase delay (θ_(y)) of the vertical supply line can be adjusted to make D equal to the angle (Ψ) of the wave tilt (W). The angle (Ψ) of the wave tilt can be determined from Equation (86). The electrical phase Φ can then be matched to the angle of the wave tilt. For example, the electrical phase delay Φ=θ_(c)+θ_(y) can be adjusted by varying the geometrical parameters of the coil L_(R) and/or the length (or height) of the vertical supply line conductor.

Next at 190, the load impedance of the charge terminal T_(R) can be tuned to resonate the equivalent image plane model of the tuned resonator 306 a. The depth (d/2) of the conducting image ground plane 139 (FIG. 9A) below the receiving structure can be determined using Equation (100) and the values of the lossy conducting medium 203 (e.g., the Earth) at the receiving structure, which can be locally measured. Using that complex depth, the phase shift (θ_(d)) between the image ground plane 139 and the physical boundary 136 (FIG. 9A) of the lossy conducting medium 203 can be determined using θ_(d)=β_(o) d/2. The impedance (Z_(in)) as seen “looking down” into the lossy conducting medium 203 can then be determined using Equation (99). This resonance relationship can be considered to maximize coupling with the guided surface waves.

Based upon the adjusted parameters of the coil L_(R) and the length of the vertical supply line conductor, the velocity factor, phase delay, and impedance of the coil L_(R) and vertical supply line can be determined. In addition, the self-capacitance (C_(R)) of the charge terminal T_(R) can be determined using, e.g., Equation (24). The propagation factor (β_(p)) of the coil L_(R) can be determined using Equation (98), and the propagation phase constant (β_(w)) for the vertical supply line can be determined using Equation (49). Using the self-capacitance and the determined values of the coil L_(R) and vertical supply line, the impedance (Z_(base)) of the tuned resonator 306 a as seen “looking up” into the coil L_(R) can be determined using Equations (101), (102), and (103).

The equivalent image plane model of FIG. 9A also applies to the tuned resonator 306 a of FIG. 18B. The tuned resonator 306 a can be tuned to resonance with respect to the complex image plane by adjusting the load impedance Z_(R) of the charge terminal T_(R) such that the reactance component X_(base) of Z_(base) cancels out the reactance component of X_(in) of Z_(in), or X_(base)+X_(in)=0. Thus, the impedance at the physical boundary 136 (FIG. 9A) “looking up” into the coil of the tuned resonator 306 a is the conjugate of the impedance at the physical boundary 136 “looking down” into the lossy conducting medium 203. The load impedance Z_(R) can be adjusted by varying the capacitance (C_(R)) of the charge terminal T_(R) without changing the electrical phase delay Φ=θ_(c)+θ_(y) seen by the charge terminal T_(R). An iterative approach may be taken to tune the load impedance Z_(R) for resonance of the equivalent image plane model with respect to the conducting image ground plane 139. In this way, the coupling of the electric field to a guided surface waveguide mode along the surface of the lossy conducting medium 203 (e.g., Earth) can be improved and/or maximized.

Referring to FIG. 19, the magnetic coil 309 comprises a receive circuit that is coupled through an impedance matching network 333 to an electrical load 336. In order to facilitate reception and/or extraction of electrical power from a guided surface wave, the magnetic coil 309 may be positioned so that the magnetic flux of the guided surface wave, H_(φ), passes through the magnetic coil 309, thereby inducing a current in the magnetic coil 309 and producing a terminal point voltage at its output terminals 330. The magnetic flux of the guided surface wave coupled to a single turn coil is expressed by

=∫∫_(A) _(CS) μ_(r)μ_(o) {right arrow over (H)}·{circumflex over (n)}dA  (104)

where

is the coupled magnetic flux, μ_(r) is the effective relative permeability of the core of the magnetic coil 309, μ_(o) is the permeability of free space, {right arrow over (H)} is the incident magnetic field strength vector, {circumflex over (n)} is a unit vector normal to the cross-sectional area of the turns, and A_(CS) is the area enclosed by each loop. For an N-turn magnetic coil 309 oriented for maximum coupling to an incident magnetic field that is uniform over the cross-sectional area of the magnetic coil 309, the open-circuit induced voltage appearing at the output terminals 330 of the magnetic coil 309 is

$\begin{matrix} {{V = {{{- N}\frac{d\; \mathcal{F}}{dt}} \approx {{- j}\; {\omega\mu}_{r}\mu_{0}{NHA}_{CS}}}},} & (105) \end{matrix}$

where the variables are defined above. The magnetic coil 309 may be tuned to the guided surface wave frequency either as a distributed resonator or with an external capacitor across its output terminals 330, as the case may be, and then impedance-matched to an external electrical load 336 through a conjugate impedance matching network 333.

Assuming that the resulting circuit presented by the magnetic coil 309 and the electrical load 336 are properly adjusted and conjugate impedance matched, via impedance matching network 333, then the current induced in the magnetic coil 309 may be employed to optimally power the electrical load 336. The receive circuit presented by the magnetic coil 309 provides an advantage in that it does not have to be physically connected to the ground.

With reference to FIGS. 18A, 18B, 18C and 19, the receive circuits presented by the linear probe 303, the mode-matched structure 306, and the magnetic coil 309 each facilitate receiving electrical power transmitted from any one of the embodiments of guided surface waveguide probes 200 described above. To this end, the energy received may be used to supply power to an electrical load 315/327/336 via a conjugate matching network as can be appreciated. This contrasts with the signals that may be received in a receiver that were transmitted in the form of a radiated electromagnetic field. Such signals have very low available power, and receivers of such signals do not load the transmitters.

It is also characteristic of the present guided surface waves generated using the guided surface waveguide probes 200 described above that the receive circuits presented by the linear probe 303, the mode-matched structure 306, and the magnetic coil 309 will load the excitation source 212 (e.g., FIGS. 3, 12 and 16) that is applied to the guided surface waveguide probe 200, thereby generating the guided surface wave to which such receive circuits are subjected. This reflects the fact that the guided surface wave generated by a given guided surface waveguide probe 200 described above comprises a transmission line mode. By way of contrast, a power source that drives a radiating antenna that generates a radiated electromagnetic wave is not loaded by the receivers, regardless of the number of receivers employed.

Thus, together one or more guided surface waveguide probes 200 and one or more receive circuits in the form of the linear probe 303, the tuned mode-matched structure 306, and/or the magnetic coil 309 can make up a wireless distribution system. Given that the distance of transmission of a guided surface wave using a guided surface waveguide probe 200 as set forth above depends upon the frequency, it is possible that wireless power distribution can be achieved across wide areas and even globally.

The conventional wireless-power transmission/distribution systems extensively investigated today include “energy harvesting” from radiation fields and also sensor coupling to inductive or reactive near-fields. In contrast, the present wireless-power system does not waste power in the form of radiation which, if not intercepted, is lost forever. Nor is the presently disclosed wireless-power system limited to extremely short ranges as with conventional mutual-reactance coupled near-field systems. The wireless-power system disclosed herein probe-couples to the novel surface-guided transmission line mode, which is equivalent to delivering power to a load by a waveguide or a load directly wired to the distant power generator. Not counting the power required to maintain transmission field strength plus that dissipated in the surface waveguide, which at extremely low frequencies is insignificant relative to the transmission losses in conventional high-tension power lines at 60 Hz, all of the generator power goes only to the desired electrical load. When the electrical load demand is terminated, the source power generation is relatively idle.

Referring next to FIGS. 20A-E, shown are examples of various schematic symbols that are used with reference to the discussion that follows. With specific reference to FIG. 20A, shown is a symbol that represents any one of the guided surface waveguide probes 200 a, 200 b, 200 c, 200 e, 200 d, or 200 f; or any variations thereof. In the following drawings and discussion, a depiction of this symbol will be referred to as a guided surface waveguide probe P. For the sake of simplicity in the following discussion, any reference to the guided surface waveguide probe P is a reference to any one of the guided surface waveguide probes 200 a, 200 b, 200 c, 200 e, 200 d, or 200 f; or variations or combinations thereof.

Similarly, with reference to FIG. 20B, shown is a symbol that represents a guided surface wave receive structure that may comprise any one of the linear probe 303 (FIG. 18A), the tuned resonator 306 (FIGS. 18B-18C), or the magnetic coil 309 (FIG. 19). In the following drawings and discussion, a depiction of this symbol will be referred to as a guided surface wave receive structure R. For the sake of simplicity in the following discussion, any reference to the guided surface wave receive structure R is a reference to any one of the linear probe 303, the tuned resonator 306, or the magnetic coil 309; or variations or combinations thereof.

Further, with reference to FIG. 20C, shown is a symbol that specifically represents the linear probe 303 (FIG. 18A). In the following drawings and discussion, a depiction of this symbol will be referred to as a guided surface wave receive structure R_(P). For the sake of simplicity in the following discussion, any reference to the guided surface wave receive structure R_(P) is a reference to the linear probe 303 or variations thereof.

Further, with reference to FIG. 20D, shown is a symbol that specifically represents the tuned resonator 306 (FIGS. 18B-18C). In the following drawings and discussion, a depiction of this symbol will be referred to as a guided surface wave receive structure R_(R). For the sake of simplicity in the following discussion, any reference to the guided surface wave receive structure R_(R) is a reference to the tuned resonator 306 or variations thereof.

Further, with reference to FIG. 20E, shown is a symbol that specifically represents the magnetic coil 309 (FIG. 19). In the following drawings and discussion, a depiction of this symbol will be referred to as a guided surface wave receive structure R_(M). For the sake of simplicity in the following discussion, any reference to the guided surface wave receive structure R_(M) is a reference to the magnetic coil 309 or variations thereof.

2. Object Identification 2(A). General Overview

With additional reference to FIGS. 21 and 22, schematically illustrated are embodiments of an object identification system 400 that uses guided surface waves as described in the preceding section to power one or more responsive tags 402. It will be re-emphasized that the appended figures are not necessarily to scale.

Each tag 402 may be associated with an object 404. The object 404 may be any type of article. Exemplary objects 404 include, but are not limited to, a consumer item, a group of goods, an article of clothing, a foodstuff, packaging for an article, a container for multiple articles, a vehicle, a pallet on which goods are stacked, a shipping container, or any other item for which tracking is desired.

The object identification system 400 includes an interrogator 406. In the embodiment of FIG. 21, the interrogator 406 includes a guided surface wave waveguide probe P and a receiver 408 that are co-located. The probe P and receiver 408 may be housed in the same structure, such as a radome, decorative enclosure, etc. In this embodiment, the interrogator 406 typically has a fixed location.

In the embodiment of FIG. 22, the probe P and the receiver 408 are not co-located. As will be described, the probe P and the receiver 408 may have a physical relationship (e.g., both may be deployed at a facility) or may have no or very little physical association. In this embodiment, the probe P and receiver 408 functionally form an interrogator 406, but are not necessarily deployed by the same party, need not be co-located, and need not be thought of as a unit. In this embodiment, the probe P typically has a fixed location and may be housed in a suitable structure such as a radome or decorative enclosure. The receiver 408 may have a fixed location or may be portable. For instance, the receiver 408 may be handheld and used by a person as the person moves about or may be mounted on a vehicle such as a truck, fork truck, aircraft, cargo ship, etc.

In the embodiments of FIGS. 21 and 22, the probe P launches a guided surface wave along an underlying terrestrial medium 410 as described in the preceding section. The terrestrial medium 410 may be any appropriate lossy conducting medium such as, but not limited to, the earth, the floor of a store, warehouse, factory or other facility, or any other appropriate substrate. As described, the probe P does not produce a radiated wave, but launches a guided surface wave along the surface of the medium 410. The energy emitted from the probe P is transmitted as Zenneck surface currents to one or more tags 402 that are located within an effective transmission range of the guided surface waveguide probe P. The probe P may be configured as any of the probes described above or in any other appropriate configuration.

With additional reference to FIG. 23, a representative tag 402 is schematically illustrated. The tag 402 is configured much like an RFID tag and includes an antenna 412 and tag circuitry 414 that are mounted to a substrate 416, such as a paper or plastic sheet. The substrate 416 may include adhesive to attach the tag 402 to the object 404. Other fastening techniques may be used or the tag 402 may form part of the object 404. In another embodiment, the tag may be located inside the object 404 or the electrical components of the tag 402 may form part of electrical components of the object 404.

In a typical embodiment, other than the drawing of power from the guided surface wave, the tag 402 does not have a power source, such as a battery or a physical connection to an external power source. Rather, the tag 402 is responsive to guided surface waves of one or more frequencies. For instance, electromagnetic energy from the guided surface wave produced by the probe P induces a current in the antenna 412 and this current is coupled to and used to power the tag circuitry 414. Similar to the way RF energy powers a conventional RFID tag, the powering of the tag circuitry 414 in this manner may be referred to as illuminating the tag 402. But, in contrast to conventional RFID tags, the tag circuitry 414 may load the probe P.

The tag circuitry 414 may include any appropriate electrical components and may be configured to carry out any appropriate functions. For example, the tag circuitry 414 may include a memory that stores data such as, but not limited to, an identifier that may be used to identify the associated object 404. The identifier may be representative of the type of goods, such as a stock-keeping unit (SKU). SKUs are unique identifiers for each distinct product available in commerce. Alternatively, the identifier may be representative of the specific item, such as a unique identifier that distinguishes the object from all other objects including objects that are nominally the same (e.g., objects having the same SKU). The tag circuitry 414 may read the identifier from the memory and, via the antenna 412 (or a second antenna, not shown) transmit an RF signal containing the identifier in a data message format. In another embodiment, the tag 402 may respond by emitting a guided surface wave, but an RF return signal is likely more convenient to generate due to a desire to keep the tags 402 relatively small, flat and power efficient.

In one embodiment, the tag is addressable and has a unique address, such as a media access control (MAC) address or an Internet protocol version 6 (IPv6) address, which includes hierarchical addressing. In one embodiment, the identifier of the tag is the same as the address of the tag.

The RF signal emitted by the tag 402 may be received by the receiver 408. The receiver 408 may analyze the signal to determine the identifier. In one embodiment, the receiver 408 communicates the identifier and any other appropriate information collected during reading of the tag 402 to a computer system 418 (FIGS. 21 and 22). For these purposes, the receiver 408 may include an antenna and radio circuit that receive the RF signal emitted by the tag 402, processing circuitry to conduct any appropriate functions connected with the reading, storing, analyzing and processing of data received from the tag 402 or ascertained at the time of reading (e.g., location data, time of arrival or signal strength as described below), and communications interfaces for establishing operative communication with the computer system 418. Therefore, the receiver 408 may include a memory for storing data and logical instructions and a processor for executing the logical instructions. Alternatively, the computer system 418 and the receiver 408 may be combined.

Upon receipt of the identifier, the computer system 418 may carry out one or more functions appropriate for the received identifier. Various exemplary functions that are carried out by the computer system 418 will be described in greater detail below.

The receiver 408 and the computer system 418 may communicate over a communications medium 420. The communications medium 420 may include one or more of a direct wired connection (e.g., a USB interface), a direct wireless connection (e.g., a Bluetooth interface), a wide area network connection (e.g., communications over the Internet) or a local area network (e.g., communications over a corporate network or WiFi network), etc. In some embodiments, the computer system 418 also may communicate with the probe P, such as to control the generation of the guided surface wave in terms of when to generate the guided surface wave, the duration of the guided surface wave production, the frequency of the guided surface wave, etc.

2(B). Powering Tags with a Guided Surface Wave

Powering RFID tags is forward link limited. More specifically, conventional RFID tags are illuminated and read by a conventional RFID interrogator (also referred to as an RFID reader). The RFID interrogator emits an RF signal using a relatively small and directional antenna. The emitted RF energy is limited, typically by a regulatory authority such as the Federal Communications Commission (FCC) in the US. The limits are present to avoid creating impermissible interference to other systems and to avoid the emission of potentially harmful radiation. Therefore, to transfer enough energy to a conventional RFID tag using conventional RFID frequencies (e.g., allocated frequencies near 900 MHz or at 13.56 MHz) to power the tag's circuitry and invoke an RF response requires close proximity between the conventional RFID interrogator and the conventional RFID tag. In most cases, the maximum distance between the RFID interrogator and the RFID tag for effective reading is a few meters, and may be shorter when the return signal from the RFID tag relies on inductive coupling with the RFID interrogator. In addition, conventional RFID technologies have poor penetration into high permittivity and lossy materials, an example of which is a pallet of water bottles or water-containing foodstuffs. Therefore, reading an RFID tag where a high permittivity and lossy material is interposed between the RFID interrogator and the RFID tag is often not successful.

Conventional RFID technologies inherently limit the functionality of the RFID tags. More specifically, there is little power available to perform processing functions, memory read operations, memory write operations, data transmit operations and so forth. At the same time, there is interest by merchants and others to extend RFID applications for inventory and supply chain control, reducing “shrinkage” of inventory caused by the theft of product, and carrying out other functions.

The techniques disclosed herein overcome these deficiencies and enhance the functions that may be carried out with tagged objects by use of a guided surface wave to supply a greater amount of power “on target” (e.g., the target being one or more tags 402). Therefore, the disclosed technique overcome the forward link limitations found with conventional RFID tags.

The tag 402 may be thought of as a load on the probe P and, in most situations, may draw as much power as needed to perform processing functions, memory read operations, memory write operations, data transmit operations and so forth. Exemplary operations will be described in greater detail below. Furthermore, the distance between the tag 402 and the probe P and the distance between the tag 402 and the receiver 408 may be greatly extended relative to the distance conventionally required between RFID interrogators and RFID tags. It is noted that the forward link to power the tags 402 in the disclosed approach may have tens of dB higher link quality than the return link between the tag 402 and receiver 408. Nevertheless, system performance will be satisfactory to carry out the functions and features described herein as well as other similar features and functions.

In order to derive power from the guided surface wave, the tag 402 includes the antenna 412. The antenna 412 may be a loop antenna (also referred to as a coil antenna) as schematically shown in FIG. 23 or may be implemented as the magnetic coil 309 schematically shown in FIG. 19. In other embodiments, the antenna 412 may be configured as a dipole antenna or as the linear probe 303 schematically shown in FIG. 18A. More than one antenna 412 may be present. In this case, the antennas 412 may be of the same type (e.g., loop antennas or dipole antennas) or may be of different types (e.g., a loop antenna and a dipole antenna). Presence of a loop antenna and a dipole antenna that are generally in the same plane or in parallel planes (e.g., both on the substrate 416) may facilitate the powering of the tag regardless of the tag's orientation. This is because at least one of the antennas will better align with the magnetic components of the guided surface wave or the electrical components of the guided surface wave, which are normal to each other. Thus, depending on spatial orientation of the tag, the loop antenna may be the dominant supplier of electrical power to the tag circuit 414 from the magnetic components of the guided surface wave or the dipole antenna may be the dominant supplier of electrical power to the tag circuit 414 from the electrical components of the guided surface wave. It is further contemplated that many conventional RFID tag antenna designs may be employed or modified to convert enough energy from the guided surface wave to electrical energy to power the tag circuit 414.

The tag circuitry 414 may include an impedance matching network as described above. In some embodiments, the impedance matching network will be statically arranged or may be omitted. A statically arranged or omitted impedance matching network may not result in maximum energy conversion performance, but will make the tag circuitry 414 relatively simple and accommodate frequent movement of the tag 402 without the need to reconfigure the impedance matching network according to its location relative to the lossy conducting medium 410. Regardless of the specific arrangement of the antenna 412, the tag 402 may be considered to include a guided surface wave receive structure R as described above in connection with FIGS. 1-20E.

The tag 402 may be relatively small and light. Most tags 402 will be similar in size and weight to conventional RFID tags. For example, the tag 402 may be relatively flat (e.g., about 1 mm thick or less), in the range of about 1 cm long to about 10 cm long, and in the range of about 1 cm wide to about 10 cm wide.

As will be described, the guided surface wave that functions as a forward link to deliver power from the probe P to the tag 402 may have one frequency and the tag 402 may emit a return link signal on a second frequency to transmit data to the receiver 408. To increase performance and data throughput for multiple tags 402 operating at low power, the second frequency may be higher than the first frequency (e.g., one or more orders of magnitude higher). To accommodate the emission of the return link signal at high frequencies, the tag 402 may include a second antenna 422 in cases where the antenna 412 is not capable of efficiently emitting the return link signal.

The system 400 may be configured to take advantage of the properties of guided surface waves as described above. Thus, practical use of a guided surface wave at a relatively low frequency may be made in connection with object identification. In one embodiment, the frequency of the guided surface wave emitted by the probe is around 13.56 MHz or other frequency that is already authorized by the appropriate regulatory authority for use with RFID technology. Frequencies higher than or lower than 13.56 MHz may be used depending on the object identification application and desired characteristics of the guided surface wave. The architecture of the tag 402, including antenna configuration and/or impedance matching, may be coordinated with the frequency of the guided surface wave to effectuate energy transfer.

As described above, field strength of the guided surface wave remains relatively high for distances from the probe P that are less than the knee 109 (FIG. 1) of the guided field strength curve 103. As such, a single probe P may be used to power many tags 200 within an effective area surrounding the probe P while maintaining an acceptable energy density at the location of the probe P. For instance, effective isotropic radiated power (EIRP) at the energy source used in connection with object identification applications may be imposed by regulatory authorities. Typical limits for conventional RFID applications are about one or two watts. One might reasonably assume that these types of EIRP limits will be maintained for some types of object identification applications using guided surface waves. Even at these limits, a single probe P may be able to power hundreds, thousands or millions of tags 402 that are located within a radial distance from the probe P that is less than the distance of the knee 109 of the corresponding guided field strength curve 103 from the probe P. For an omnidirectional probe P, the effective area in which tags may be illuminated is a circular area have a radius that is about the distance of the knee 109 of the corresponding guided field strength curve 103 from the probe P. The distance of the knee 109 from the probe P is dependent on frequency of the guided surface wave. As an example, the distance of the knee 109 from the probe P for a guided surface wave at about 13 MHz is approximately one kilometer, depending on ground properties. Under relatively ideal circumstances, conventional RFID technology operating at 900 MHz has an effective operating range of about 30 meters. Therefore, it will be appreciated that tags 402 may be powered from a much greater distance and at a much lower frequency than previously possible.

A tag 402 may be configured to respond (e.g., become powered and/or transmit a return signal) when illuminated with a guided surface wave of a predetermined frequency, multiple frequencies or a range of frequencies. In one embodiment, a tag 402 is configured to respond to a first frequency, but not a second frequency, and a different tag 402 is configured to respond to the second frequency, but not the first frequency. In one embodiment, a minimum separation between the first and second frequencies may be established, such as about a 10 kHz separation or a 100 kHz separation.

As will be appreciated, many tags 402 may be powered efficiently with the use of a guided surface wave and the tags 402 may be configured to carry out relatively power intensive functions. Many of these functions will be described below. Moreover, the use of inductive readers with limited operational range may be avoided. This allows for the interrogation of tags 402 at a significant distance and/or with relatively low frequencies. The nature of guide surface waves also allows for the interrogation of a tag 402 in situations where high permittivity material and/or lossy material is interposed between the probe P and the tag 402. As an example, a tag 402 located within a pallet of goods or a shipping container of goods having water content (e.g., water-containing foodstuffs such as water bottles, beer, soup, condiments such as ketchup or barbeque sauce, etc.) may be interrogated. In one embodiment, a tag 402 may be powered to operate when there is one to five meters of water interposed between the probe P and the tag 402.

2(C). Tag Interrogation

One or more tags 402 may be interrogated (also referred to as read) by illuminating the tag 402 with a guided surface wave having a frequency compatible with the tag 402 and receiving a return signal from the tag 402 with a receiver 408. As part of this process, the tag 402 draws power from the guided surface wave to power the electronics (tag circuit 414) in the tag 402. The drawing of power may be a passive operation. Specifically, the guided surface wave induces a current in the antenna 412 that is applied to the tag circuitry 414. The application of power to the tag circuitry 414 activates the tag circuitry 414 to carry out one or more predetermined functions. An exemplary predetermined function is to read the tag identifier associated with the tag 414 from a memory component of the tag circuit 414 and transmit a return signal containing the tag identifier. The return signal may be in the form of a data transmission that follows a predetermined protocol in terms of time of transmission (e.g., a predetermined time slot under time division multiplexing with the return signals of other tags), electric characteristics, message format or content, encryption, etc. The signal may be received by a receiver 408 and interpreted.

In one embodiment, the return signal may be an RF signal. The propagation capabilities of the return signal will depend on the characteristics of the RF signal, such as energy level, data encoding and frequency. The distance at which the return signal may be effectively detected by a receiver 408 will depend on the propagation capabilities of the return signal in the surrounding environment and the sensitivity of the receiver 408. To enable reading at relatively large distances, such as greater than 30 meters, the return signal may be emitted with a relatively large EIRP. Drawing power from a guided surface wave will allow a transmitter in the tag circuit 414 to radiate with relatively high power since the energy density of power available in the source (the guided surface wave) is high. Additionally, the return signal may have a frequency that is relatively high to enhance throughput. In one embodiment, the return signal may have a frequency higher than the frequency of the illuminating guided surface wave, such as about one to three orders of magnitude higher than the frequency of the guided surface wave. For example, if the guided surface wave is in the range of about 10 MHz to about 250 MHz, the return signal may be in the range of about 100 MHz to about 5.4 GHz, or higher.

Therefore, the guided surface wave may be at one frequency (e.g., a first frequency) and the tag 402 may respond at a second frequency that is different than the first frequency. In other embodiments, the response frequency may be nominally the same as the guided surface wave frequency. In one embodiment, a first set of tags 402 responsive to a guided surface wave at a first frequency may respond at a second frequency and a second set of tags 402 responsive to a guided surface wave at the first frequency may respond at a third frequency different than the second frequency. Using the difference in response frequencies, the tags of the first set may be distinguished from the tags of the second set.

As indicated, one or more predetermined functions may be carried out by the tag 402 when the tag circuitry 414 becomes activated. One exemplary predetermined function is to emit a return signal. The return signal may contain information, such as one or more of an indication that the tag 402 is present with no identifying information, an indication of the type of tag 402 or the type of object 404 with which the tag 402 is associated, a SKU or other identifier for the object 404 with which the tag 402 is associated, a unique identifier or address of the tag 402 that distinguishes the tag 402 from other sets of tags 402 or from all other tags 402, or any other data stored by the tag 402.

In one embodiment, the transmission of the return signal is automatic. In other embodiments, the response or other action taken by the tag 402 may be carried out under certain conditions. In an exemplary embodiment, the tag 402 is addressable and responds to messages or data addressed to the tag 402. Depending on the addressing scheme, the tag 402 may be individually addressable. For this purpose, the tag 402 may have any address that is unique from the addresses of all other tags 402, such as an IPv6 address or some other address of appropriate format. In one embodiment, the address may have a length that is about 40 bits to about 64 bits. It is contemplated that addresses that are 64 bits long or longer may be used to uniquely address every object on the planet. In other embodiments, a message or command may be addressed to plural tags 402. For this purpose, tags 402 may share a common address (e.g., all tags 402 associated with a SKU may have the same address) or hierarchical addressing may be used to take advantage of otherwise unique addresses. Other exemplary data distribution techniques include multicast addressing or geocasting.

Using addressable tags 402 allows for various predetermined functions to be carried out by the tags 402. As an example, a data link or communications interface (e.g., Bluetooth interface) between the receiver 408 and the tag 402 may be established for the bidirectional exchange of data. Communication between the receiver 408 and the tag 402 may allow the receiver 408 (or the computer system 418 via the receiver 408) to poll the tag 402 for information stored by the tag 402 or send commands to the tag, or may allow the tag 402 to receive and store additional information.

In another embodiment, the predetermined function that is carried out by the tag 402 includes storing data encoded in the guided surface wave or carrying out a command that is encoded in the guided surface wave. The data or command in the guided surface wave to which the tag 402 is responsive may be broadcast to tags 402 without addressing or may be addressed to one or more specific tags 402. For this purpose, the probe P may include an encoded carrier message in the guided surface wave.

Predetermined functions that may be carried out by one or more tags 402 when data and/or commands are transmitted by the receiver 408 or as part of the guided surface wave may include, but are not limited to, writing data to a memory of the tag 402, executing a command, responding with requested information, and responding by emitting a return signal only if addressed or otherwise polled.

Another predetermined function may be to stop emitting the return signal in response to a message acknowledging receipt of the return signal or an appropriate command. This function may be employed in various situations. For instance, during an inventory control operation, the guided surface wave may be used to illuminate many tags 402, all of which may commence response operations by emitting respective return signals. As responses from individual tags 402 are received and processed, the computer system 418 may issue commands (via the receiver 408 or the guided surface wave) to the tags 402 from which return signals are received and processed to stop emitting return signals. In this manner, the return signals from other tags 402 may be received and processed with less contention.

In one embodiment, it may be possible to permanently “turn off” or deactivate a tag 402 by executing a command in the tag 402. For instance, after an object 404 is purchased by a consumer, its associated tag 402 may be deactivated so that the tag will no longer carry out predetermined functions when illuminated by an appropriate guided surface wave.

2(D). Regionalizing Tag Illumination

Additional reference is made to FIG. 24. FIG. 24 shows two adjacent sites 424 a and 424 b. The sites 424 in the illustrated embodiment are buildings that each house a retail establishment. This exemplary embodiment is shown for descriptive purposes. It will be appreciated that the illustrated embodiment is representative of aspects of the disclosed concepts. The nature and configuration of sites at which principles of the disclosed concepts are applicable may vary. Types of sites include, but are not limited to, retail establishments, warehouses, office facilities, schools, ports, fulfilment centers, shipping and sorting centers, sporting venues, parking lots, factory or manufacturing establishments, farms, military bases, etc. The sites may not include any building structures or may include one or more building structures. Each site is characterized by a known geographical area in which the illumination and reading of tags 402 is desired. Due to the relative size of the tags 402 and the sites, individual tags 402 and associated objects 404 are not shown in FIG. 24 for simplicity of illustration. But it will be understood that tags 402 and associated objects 404 are present within each site 424. The number of tags 402 and associated objects 404 in a site 424 may vary and could range from as little as one tag 402/associated object 404 to millions of tags 402/associated objects 404.

In the illustrated embodiment, the sites 424 a and 424 b are spaced apart. Adjacent sites 424 need not be spaced apart. Sites 424 that correspond to buildings may touch or nearly touch one another, or may share a wall that demarks one site 424 from another.

In one embodiment, a probe P is associated with each site 424. Typically, the probe P is located within the geographic area that defines the site 424. One or more receivers 408 are also associated with and located at the site 424. Typically, the receivers 408 that are associated with a site 424 are located within the geographic area that defines the site 424, but one or more of the receivers 408 associated with a site 424 may be located outside this geographic area such as near an entrance to the site 424.

Each probe P is configured to illuminate tags 402 located within the geographic area of the site 424 associated with the probe P. In one embodiment, the probe P associated with one site 424 is configured to not illuminate tags 402 that are located within an adjacent site 424. It will be appreciated that not illuminating tags 402 in an adjacent site may not always be possible or practical, and/or sometimes tags 402 in an adjacent site may be inadvertently illuminated even if care is taken to limit the operable range of a probe P.

For the purpose of configuring a probe P to not illuminate tags 402 in an adjacent site, the natural “energy bubble” resulting from the generation of the guided surface wave by the probe P may be employed. As described above, the energy density roll-off of a guided surface wave is very low at distances less than the distance of the knee 109 from the probe P. At distances at the knee 109 and outward, the energy density falls of dramatically. The energy density behaves in this manner in all radial directions from the probe P assuming that the probe P is omnidirectional and the electrical properties of the terrestrial medium 410 are uniform along the operative interface between the probe P and the terrestrial medium 410. The distance of the knee 109 is a function of the frequency of the guided surface wave. Also, for purposes of this description, to be considered illuminated, a tag 402 must be in the presence of a threshold energy density to draw sufficient power from the guided surface wave to be powered on and capable of responding. The threshold energy density may depend on the energy consumption characteristics of the tag 402 and, therefore, may vary.

For tags 402 that are operatively compatible with the frequency of a guided surface wave generated by a probe P, the area surrounding the probe P in which the tags 402 will be exposed to the threshold energy density to become illuminated will be referred to as an illumination area 426. As illustrated in the exemplary embodiment of FIG. 24, there is one illumination area 426 a associated with the site 424 a and probe Pa and another illumination area 426 b associated with the site 424 b and probe Pb. In an embodiment where the frequency of the guided surface waves generated by the probes Pa and Pb for neighboring sites 424 a, 424 b are operatively compatible with the tags 402 used in the other of the neighboring sites 424 a, 424 b, the establishment of non-overlapping illumination areas 426 will allow for each site 424 to conduct object identification by reading tags 402 independently of one another.

In particular, the guided surface wave generated by the probe Pa for site 424 a will tend not to illuminate tags within the neighboring site 424 b and vice versa. Additional precautions may be taken to avoid having receivers 408 for one site 424 a detect the responsive signals from tags 402 located in the neighboring site 424 b when the tags 402 are illuminated by the probe Pb for the neighboring site 424 b and vice versa. These precautions may include controlling the timing of illumination so that the probes Pa, Pb from the respective sites 424 a, 424 b are not actively generating guide surface waves at the same time. Another precaution is limiting the output power of the tags 402 to a level low enough to avoid detection by receivers 408 in the other site and/or limiting the receive sensitivity of the receivers 408 to avoid detection of signals from tags 402 in the neighboring site 424. Another precaution is maintaining, in a computer system 418 that processes information from the readers 408 of a site 424, a database of the tag identifiers for all the tags 402 that should be present in the site 424. If a tag 402 is read and the associated tag identifier is not in the database, an assumption may be made that the tag 402 is not associated with the site 424 and should be ignored. An exception may be made in an intake mode when objects arrive at the site 424 and are interrogated to add the corresponding tag identifiers to the database.

Noting the foregoing, there are several factors that control the effective size of the illumination area 426, including power and frequency of the guided surface wave and the power requirements of the tag 402. Therefore, each of power and frequency of the guided surface wave, the characteristics of the tags 402 used within the site 424, and the characteristics of the tags 402 used in neighboring site(s) may be selected in coordination with each other to establish an appropriate size for each illumination area 426. It will be appreciated, however, that frequency is the most significant contributing factor to size of the illumination area 426. A frequency in the range of about 100 MHz to about 200 MHz should be sufficient to control the size of the illumination area 426 to closely match the size of the site 424 when the site 424 is a typical warehouse or retail establishment.

It also may be desirable to control the shape of the illumination area 426. Shape of the illumination area 426 may be controlled by using a probe assembly with output that varies as a function of direction. This may be achieved using plural probes P to create lobes in the guided surface wave profile or create a guided surface wave that is the aggregate of plural directionally launched guided surface waves (e.g., a multi-beam approach). For instance, super-positioning of individual probes P may be used to make a phased array probe with directional output that is controlled by the presence of multiple, simultaneously generated guided surface waves.

By selecting characteristics of the probe P (or probe assembly) and tags 402 to control the size and shape of the illumination area 426, the illumination area 426 may be made to approximate the geographic area of the associated site 424. Also, as described above, it may be possible to use one probe P in the site 424 to illuminate all tags 402 in the site 424 (e.g., by achieving high energy density across the entire site 424) while maintaining an acceptable energy density at the source (e.g., an EIRP of about 1 watt to about 2 watts at the probe P).

Additional considerations may be used in selecting the frequency of the guided surface wave. For instance, access to certain frequencies may or may not be made available for object identification purposes by the regulatory authority overseeing the jurisdiction in which the site 424 is located.

Another consideration is effective height of the guided surface wave. Energy density of a guided surface wave falls off at a height of about a wavelength of the guided surface wave. Therefore, the height of the illumination area 426 will be about a wavelength of the guided surface wave. For a guided surface wave of about 13 MHz, the probe P will be about three feet tall and the illumination area 426 will be about 72 feet tall (about 22 meters). This height may be sufficient to illuminate tags 402 associated with objects 404 that are placed on upper shelves in many warehouses. For a guided surface wave of about 100 MHz, the illumination area 426 will be about 3 meters tall and, for a guided surface wave of about 300 MHz, the illumination area 426 will be about 1 meter tall. These heights may be compatible with many retail environments.

2(E). Data Collect from Tags at a Site

Various functions may be carried out by reading tags 402 that are present at a site. Exemplary functions include inventory control, finding misplaced objects 404, reducing theft, and consumer transaction operations. For these tasks, it will be assumed that each object 404 to be tracked is associated with a tag 402 and the computer system 418 maintains a database of the objects 404 and each associated tag identifier. This information may be generated and/or gathered when the tag 402 is first associated with the object 404, which may occur at a location remote from the site 424 such as at a factory where the object is manufactured. In other situations, this information may be generated and/or gathered when the tag 402 arrives at the site.

To carry out reading of tags 402 at the site 424, one or more probes P and one or more receivers 408 are present. Since tags 402 may be illuminated by a guided surface wave generated by a probe P that is located outside the site 424, the probe P need not be located within the geographic area of the site 424. But it is contemplated that each receiver 408 that receives return signals from tags 402 located at the site 424 will be located in the geographic area of the site 424 or close to the site 424 (e.g., within a distance capable of receiving return signals emitted by tags 402 in the site 424).

Each receiver 408 for a site 424 may be strategically placed, such as by doors, loading docks, cash registers, etc. For example, in the illustrated embodiment of site 424 a where site 424 a is a retail location, a receiver 408 is located adjacent a main entrance 428 though which customers enter and exit, a receiver 408 is located adjacent a door 430 that separates a main shopping area 432 from an inventory storage area 434, and a receiver 408 is located adjacent an ancillary exit door 436 at the storage area 434. Another receiver 408 may be located adjacent a loading dock 438 and another receiver 408 may be located at a payment area 440. Objects 404 and associated tags 402 may be present on shelves 442 or displays located in the shopping area 432. Additional objects 404 and associated tags 402 may be present on shelves 442 or in other locations in the storage area 434. Receivers 408 at additional or alternative locations also may be present.

With additional reference to the illustration of the exemplary site 424 b in FIG. 24, another arrangement for the receivers 408 will be described. In this embodiment, receivers 408 are placed at strategic locations but are not associated with specific locales within the site 424 b such as doors, loading docks, payment areas, etc. Rather, the receivers 408 are positioned to detect return signals emitted by tags 402 that are within the site 424. Although two receivers 408 are illustrated in the appended figure, other numbers of receivers 408 are possible. For example, there may be only one receiver 408 or three or more receivers 408. The return signals may be used and analyzed in the same manners as described above. The number and positioning of receivers 408 in either the embodiment of site 424 a or site 424 b may depend on the operative range between tags 402 and receivers 408, the size of the site 424, programming of the computer system 418 and any other relevant factors. Also, the receiver arrangement of the embodiment of site 424 a may be combined with the receiver arrangement of the embodiment of site 424 b so that some receivers are positioned in connection with certain structural elements of the site and others are positioned at more generic strategic locations.

It will be recognized that the locations of the receivers 408 in FIG. 24 are exemplary and for descriptive purposes. The number and location of receivers 408 may be modified depending on the characteristics of the site 424 and tag reading functions to be performed.

The probe P may be located in a strategic location, but may be hidden from sight. For example, in the embodiment of site 424 a, the probe Pa is hidden in an end cap 444 of one of the shelves 442. The probe P may be configured to generate the guided surface wave continuously so that each tags 402 in the respective illumination region 426 responds continuously, such as by retransmitting the return signal without delay between retransmissions or periodically retransmitting the return signal (e.g., once a second). In other embodiments, the probe P is controlled to generate the guided surface wave at desired times and for desired durations. The desired times may be prescheduled or may be the result of triggering the activation of the probe (e.g., an operator may trigger the probe to conduct an inventory check, to find a misplaced object or to tally objects for purchase as described in the following exemplary functions).

The return signals may be detected by one or more receivers 408. Data derived from the return signals (e.g., the tag identifiers), together with the known locations and/or identities of the receivers 408 that detect the return signals, may be used in connection with various functions. One exemplary function is to assist in identifying objects 404 that a customer intends to purchase. For instance, a customer may bring objects 404 for purchase to the payment area 440. In the embodiment of site 424 a, the objects 404 may be moved passed the receiver 408 at the payment area 440 and those objects 404 may be logged by the computer system 418. It is noted that there is no need for the items for purchase to be read one at a time in similar manner to the way printed SKUs are serially scanned with a bar code reader. Rather multiple objects 404 may be brought past the receiver 408 at the same time. Once the objects 404 are identified, the customer may then pay for the items in the conventional manner.

In another embodiment, information about inventory at the site 424 may be tracked. For example, in the embodiment of site 424 a, as objects 404 enter or leave the site 424, the associated tags 402 may pass by one of the receivers 408 located at the door 428, the door 436 or the door 438. By keeping track of the objects 404 that pass these receivers 408, an accurate tally of the number of objects 404 by object type may be made and detection of the object moving from an authorized area to an unauthorized area made be made. This detection also may be made by detecting movement past a predetermined point or crossing a boundary between an authorized area and the unauthorized area. In another embodiment, detection that an object has left an authorized area may be made by failing to receive a return signal from the associated tag within a predetermined amount of time since the receipt of a last iteration of the return signal. Also, this information may be cross referenced against valid object purchases and other valid reasons why an object may be removed from the site 424 (e.g., shipped to a downstream location in a supply chain or returned to a supplier). If the departure of an object 404 is not associated with a valid reason, then additional security-related actions may be carried out, such as alerting an authority (e.g., a manager of the site 424 or the police), turning on a security camera and recording video of the area surrounding the door or dock through which the object exited, launching an investigation, etc.

Other information may be determined from the manner in which an object 404 enters or exits the site 424, the time of receipt of the return signal, and/or additional information such as when a particular vehicle or worker was also present. For example, in a facility with multiple loading docks, tracking the dock through which an object moves may be used to establish which employee handled an object, which truck an object was loaded onto or which truck brought an object to the facility. As another example, tracking of objects 404 located in the storage area 434 versus the shopping area 432 may be made by the receipt of return signals by the receiver 408 at the door 430. Other data collection regarding movement of objects within the site 424 may be made, such as tracking movement from one user-defined zone to another user-defined zone, collecting data regarding the behavior of customers, etc.

In another embodiment, an inventory of all objects 404 or certain categories of objects in the site 424 may be made by analyzing return signals from the tags 402. In one embodiment, the computer system 418 may analyze the tag identifier associated with each distinct return signal to conduct an inventory analysis. In one embodiment, de-interleaving techniques may be applied to ignore or turn off return signals from tags 402 having associated tag identifiers that have been logged into the inventory analysis. To limit the number of tags 402 that respond during an inventory analysis, addressed commands to emit a return signal may be sent to the specific tags 402 of interest. De-interleaving and/or addressing tags to respond or not respond may be used in conjunction with the other functions described herein.

In one embodiment, the geo-location of all objects 404, certain categories of objects 404 or a single specific object 404 may be identified using the return signals from the tags 402 associated with the objects 404. The location of a tag 402 and its associated object 404 may be determined by illuminating the tag 402 and receiving the return signal at two or more receivers 408 that each have a known location. For two or more return signals from the same tag 402, time difference of arrival or differences in received power (e.g., voltage standing wave ratio or VSWR) may be used to triangulate the location of the tag 402. This analysis may be repeated for the return signals received from multiple tags 402. Also, de-interleaving techniques may be applied to ignore or turn off return signals from tags 402 for which locations have been determined. Also, to limit the number of tags 402 that respond during a location analysis, addressing may be used to control which tag or tags 402 emit a return signal.

A location determination technique (e.g., the foregoing triangulation techniques) may be used in conjunction with various functions. For instance, with reference to the exemplary depiction of site 424 b, bulk identification of objects 404 in a particular area may be made. For example, there may be a reading zone 446 that serves as a designated interrogation area through which items for purchase travel before exiting the site 424 b. All of the objects 404 in the dedicated reading zone 446 may be detected by analyzing return signals from tags 402 located in the reading zone 446. Therefore, a group of objects may be moved through the reading zone 446, collectively identified and logged by the computer system 418. Then a transaction may be completed to purchase the items. This approach to bulk object identification may be applied in other contexts, such as identification of all items moving through a loading dock, identification of all items on a truck or rail car as the truck or rail car moves through a predetermined area, etc.

As another example, geo-location may be used to detect unauthorized movement of an object 404 (e.g., theft of the object 404). In one embodiment, this detection may be made if an object 404 is determined to be in a location where it should not be present (e.g., the location of the object 404 is detected to be outside the geographic area of the site 424). In another embodiment, this detection may be made if an object moves more than a threshold distance and in an unauthorized direction from a predetermined point. This technique may detect an object moving away from a door and toward a parking lot, for example. Once a detection of possible unauthorized movement is made, the detection may be cross referenced against any legitimate reasons for the movement such as purchase of the object, scheduled shipping of the object to another location, etc. If no legitimate reason is present for the detection having been made, then security measures may be triggered. The security measures may include, but are not limited to, alerting an authority (e.g., a manager of the site 424 or the police), turning on a security camera and recording video of the area surrounding the door or dock through which the object exited, launching an investigation, etc.

In another embodiment of determining geo-location of an object 404, the geo-location of the receiver 408 may be used as a proxy for the location of the object 404 for which an associated tag return signal is received. For example, if the receiver 408 at the payment area 440 detects the return signal for a tag 402, the associated object 404 will be assumed to be located at or near the payment area 440. In the event that more than one receiver 408 detects the return signal for a tag 402, then the location of the receiver 408 that detects the highest signal strength for the return signal may be used as a proxy for the location of the associated object. In some embodiments, the receiver 408 may be mobile, such as a receiver 408 that is mounted on a truck, ship, train or other vehicle. In this case, the geo-location of the receiver 408 that serves as a proxy for the geo-location of the tag 402/object 404 may be determined using, for example, global positioning system (GPS) technology.

Any of the foregoing approaches for determining the geo-location of the tag(s) 402 may include the determination of the elevation of the tag(s) 402 in addition to geo-location (e.g., as expressed by two dimensional coordinates). Also, in some embodiments, it may be possible to refine the locating of tags 402 by steering the guided surface wave such that the guided surface wave only illuminates tags 402 in certain areas of the site 424 at a time (e.g., by using a multi-beam guided surface wave generation approach to output a guided surface wave that changes in direction over time).

Storing objects in a facility (e.g., warehouse, fulfillment center, storage area of a retail store, etc.) typically involves detailed planning of where objects are to be placed so that they may be readily found when desired. Using the disclosed techniques for illuminating and geo-locating tags 402, less planning may be employed. Instead, objects 404 may be placed in any location that will accommodate the objects 404. This location may be determined at the time of placement using one of the foregoing approaches for determining the geo-location of the tag(s) 402 that are associated with the objects 404. This location may be stored in a database by the computer system 418 and used to facilitate retrieval of the objects 404 at a later time. Alternatively, the objects 404 may be placed in a suitable location without determining or storing information about the location. When the objects are desired to be found, one of the foregoing approaches for determining the geo-location of the tag(s) 402 that are associated with the objects 404 may be used to determine the location of the objects 404.

In one embodiment, the movement of an object 404 may be tracked by periodically or continually making location determinations of the geo-location of the tag 402 that is associated with the object 404. Movement tracking in this manner may be used for inventory planning, for monitoring for theft or product shrinkage, and for a variety of other purposes. In one embodiment, the tracking of plural tags 402 may provide additional information. For instance, if a person is associated with a first tag 402 and an object 404 is associated with a second tag 402 and the tags are found to move together, a determination may be made that the person is moving the object or is associated with the movement of the object (e.g., both are moving together in a vehicle). The same analysis may be made for tags 402 associated with vehicles and tags associated with objects 404.

A tag 402 may be associated with a person in a number of manners and for a variety of purposes. In one embodiment, a tag 402 that is associated with a person may take the form of, or is included in, an object regularly carried by the person, such as a tag 402 that is similar in form factor to a credit card or a tag 402 that is part of an electronic device (e.g., mobile phone or case therefor). Once a tag 402 is associated with a person, identifying the tag, and hence the person, may be used for a variety purposes. For instance, a tag 402 that is associated with a person may be detected at the payment area 440 in connection with the detection of tags 402 associated with objects that the person intends to purchase. If a bank account, credit card or other payment means is further associated with the tag 402 that is associated with the purchasing person, then payment for the objects 404 may be made by the computer system 418 registering a transaction using the payment means that is associated with the tag 402 that is associated with the purchasing person.

In another embodiment, employees at the site 424 may be required to carry a tag 402. Using location tracking and/or associations of objects 404 with the person, a variety of functions may be carried out by the computer system 418. Exemplary functions may include tracking task completion, tracking job performance, tracking worked hours, and monitoring for theft of objects 404 by the employees.

2(F). Macro Illumination of Tags

The previous section described the use of guided surface waves to illuminate tags 402 in a well-defined geographic area corresponding to a known venue that is typically controlled by one party.

Another embodiment will be described in connection with FIG. 25. In this embodiment, a guided surface wave may be used to illuminate tags 402 over areas in which there may be multiple sites 424, over areas in which multiple receivers 408 controlled by respective parties are present, and/or over areas in which tags 402 may travel by vehicle (e.g., truck, car, plane, train, ship, etc.). The areas may include arbitrary areas, paths along which goods are intended to travel, postal codes, cities, counties, states or provinces, countries, continents, or an area determined by the operator of the probe P that may or may not correspond to regulatory boundaries, governmental boundaries or geographic boundaries. In one embodiment, the guided surface wave may be produced to illuminate tags 402 on a global basis (i.e., world-wide). Due to the size of the tags 402 and receivers 408 relative to the size of some of the contemplated areas, individual tags 402 and receivers 408 are not shown in FIG. 25 for simplicity of illustration.

Noting that the probe P is not drawn to scale and may be located almost anywhere on the planet, the representative embodiment illustrated in FIG. 25 contemplates a guided surface wave that is capable of illuminating tags 402 on a global basis. Aspects of the following description, however, also will apply to a smaller illuminated area.

The guided surface wave preferably has a known, fixed frequency (e.g., a first frequency). One or more additional probes P may be used to generate a guided surface wave(s) that illuminates tags in at least an area overlapping the area in which tags 402 are illuminated by the guided surface wave of the first frequency. The other guided surface wave(s) may have a frequency different than the first frequency and functions carried out in connection with the illumination of tags 402 with the other guided surface wave(s) may be the same or similar to the functions carried out in connection with the illumination of tags 402 with the guided surface wave of the first frequency. Therefore, the illumination of tags 402 over relatively wide-spread areas will be described in the context of a single guided surface wave of the first frequency and further in the context of tags 402 that are operationally compatible with the first frequency (e.g., are powered by the guided surface wave of the first frequency and are capable of emitting a return signal when powered on). The operation of guided surface waves of other frequencies and tags that are compatible with those other frequencies may be carried out in the same manner and in parallel with the operation of the guided surface wave of the first frequency and tags compatible with the first frequency.

In general, as the area in which tags 402 may be powered by the guided surface wave of the first frequency increases, the first frequency will decrease.

Entities that are interested in using the guided surface wave of the first frequency and generated by the probe P to power tags 402 may deploy tags 402 that are compatible with the first frequency. Deploying tags 402 may include, for example, physically associating a compatible tag 402 to each object 404 that the entity wishes to track and logging the identity of the object 404 and associated tag identifier in an appropriate database at a computer system 418 (shown not to scale). Physically associating a tag 402 and an object 404 may include adhering or securing the tag 402 directly to the object 404, to the packaging for the object 404 or some other item that is retained with the object 404 (e.g., a manual). In other embodiments, the tag 402 may be inside the object 404 or an integral part of the object 404.

The entities also may deploy receivers 408 in strategic locations in the area in which the guided surface wave will illuminate the tags 402. In addition to or instead of deploying its own receivers, an entity may cooperate with another party that deploys receivers. The other party may provide information (e.g., tag identifiers) present in return signals detected by receivers to the entity. The providing of information may be through the computer system 418 and may include processing the data to make various determinations, such as route tracking. It will further be appreciated that there may be multiple computer systems 418 that process information from return signals. For example, each entity that is interested in using the guided surface wave of the first frequency to identify objects may deploy a computer system 418 or multiple computer systems 418 to process information for multiple sites.

It is contemplated that wide-area illumination of tags 402 will lead to a number of object identification and tracking functions that are not currently possible with conventional RFID technology. In addition, any of the operations carried out when using a local probe P (e.g., as described in connection with the embodiments of FIG. 24) also may be carried out using a remote probe P as described in connection with FIG. 25.

Similar to the operations described above, tags 402 that are illuminated with the guided surface wave will respond with an identifier. The identifier may be a unique identifier to distinguish the tag 402 from all other tags 402, such as an IPv6 address or identifier in another format. The guided surface wave of the first frequency has enough energy density over the covered area, which may be up to the entire planet, to illuminate all tags 402 within the covered area. As a result, the tags 402 may continually re-radiate by emitting its return signal, which is typically done at a second frequency higher than the first frequency. Continually re-radiating a return signal may include repeating the return signal with no delay or a slight delay (e.g., up to five seconds in one embodiment, up to two seconds in another embodiment, up to one second in another embodiment, or up to 0.5 seconds in another embodiment) between return signal emissions. In some situations, tags 402 may be programmed to respond at certain times, with certain periodicity, or in response to a command to respond. In other situations, tags 402 may be commanded not to respond at least for a specified period of time (e.g., during a read operation of plural tags that employs a de-interleaving approach to accurately identify large numbers of tags).

In one embodiment, as long as a tag 402 is in the area illuminated by the guided surface wave of the first frequency, the tag 402 will radiate its identifier “all the time” (e.g., repeatedly radiate the identifier over and over again with no or little delay between each radiation cycle) and for the life-cycle of the tag 402. As such, the tag 402 may be tracked anywhere in the area illuminated by the guided surface wave of the first frequency as long as the tag 402 is within operative range of a receiver 408 that is configured to detect return signals on the emission frequency of the tag 402 (e.g., the second frequency). As previously described, the location (e.g., longitude and latitude) and elevation of a tag 402 may be determined using, for example, triangulation or by using a receiver's location as a proxy for the tag's location.

In the exemplary embodiment where the covered area is the entire world, each compatible tag 402 may be tracked anywhere on the planet at any time until the tag 402 stops transmitting. The tag 402 may stop transmitting by being deactivated in response to a deactivation command, by failure of the tag circuitry 414, by becoming physically damaged, etc. In the global embodiment, the guided surface wave may be operative to illuminate tags 402 at a relatively high altitude, such as up to about 35,000 feet. As such, tags 402 carried by an aircraft may be tracked provided a receiver can detect the reply signals from the tags 402.

Receivers 408 may be positioned at any location where tag 402 identification is desired. A non-exhaustive list of possible locations for receivers 408 includes manufacturing facilities, farms, warehouses, fulfillment centers that process Internet orders or mail orders for goods, retail locations, restaurants, grocers, ports of entry for a country, seaports, airports, along roadways, along railroad tracks, and on moving vehicles (e.g., cars, trucks, planes, ships, trains, fork trucks, etc.).

The widespread deployment of receivers 408 may allow for lifetime tracking of an object 404 that is associated with a tag 402. The amount of tracking information that is collected may depend on, for example, the nature of the object 404 associated with the tag 402, a supply chain of interest, or the interest level of persons or entities that have a relationship to the object. As an example, an object 404 may be associated with a tag 402 at the time of manufacture or packaging in a factory in Beijing, China and then tracked when loaded on a truck and driven to a seaport in Tianjin, China. Next, the object 404 is tracked when it is loaded on a cargo container and tracked when the cargo container is loaded onto a ship. The object 404 may be further tracked in route by the ship to a seaport in Los Angeles, Calif., U.S. The unloading of the cargo container from the ship and the subsequent loading of the object 404 on a train may be tracked at the seaport by receipt of the return signals from the tag 402. The object 404 may be tracked during travel by train, which may take the object to Memphis, Tenn., U.S. where it is unloaded from the train and transported to a shelf in a fulfillment center in Memphis. An order for the object from a customer in Boston, Mass., U.S. may be received by the operator of the fulfillment center. At that point, the object 404 may be removed from the shelf, placed in a shipping box, transported to a package delivery carrier's Memphis sorting and distribution center where the box containing the object is ultimately loaded on a plane. All of those events also may be tracked. The object may be tracked as the plane travels to Boston. Then tracked are events such as the offloading of the object from the plane, the transportation of the object to the package delivery carrier's Boston sorting and distribution center, the loading of the object on a delivery truck and ultimate delivery to the customer workplace or residence. Later, the customer may travel with the object 404 on vacation to Paris, France. Assuming that the associated tag 402 is not separated from the object or disabled, the object may be detected again during travel to, or while in, Paris.

It will be recognized that the foregoing object lifecycle tracking example describes a representative supply chain situation. Objects that are tracked using tags 402 that are responsive to guided surface waves may enter and pass through commerce in many other ways, but still may be tracked for a variety of purposes. Those purposes include, for example, supply chain management, inventory management, detecting theft, estimating time of arrival at a location, etc.

Detailed information regarding where an object has been, and/or persons or entities that have interacted with the object, may be used in a number of contexts. As an example, the identity of a purchaser of the object may be determined together with the vendor of the object, the retail location (if applicable) and the manner of payment (e.g., including a specific credit card, if applicable). This information may be combined and analyzed with other information about the purchaser to generate marketing opportunities, to automatically register the product for warranty, for follow-up service/product update purposes, or for other reasons.

In one embodiment, the disclosed identification and tracking technique may be used to trace the origins of a breakout of a food-borne illness. In this embodiment, the stricken persons may be interviewed to determine what the people ate, when they ate those items, and the source of the food to the person (e.g., the restaurant at which food was consumed or the grocery at which food was purchased). The information for each affected person may be populated into a database and crossed referenced to determined which food item most likely caused the illness. Sometimes merely cross referencing this information may not be sufficient to determine the food that contains a pathogen, especially if the food is distributed across wide areas of a country or region. Using information collected from the tags 402 associated with objects in the food supply chain may be of use in discovering which food is making people sick, where that food came from and where in the distribution chain other potentially contaminated food is currently located.

For this purpose, tags 402 may be associated with food items as early as possible in the food chain. For instance, tags 402 may be associated with jars of peanut butter or boxes of multiple jars of peanut butter at the processing plant that manufacturers the peanut butter and/or fills the jars. Produce (e.g., fruits and vegetables) may be associated with tags 402 at the grower or a packing facility that packages the produce (e.g., typically by placing the produce in containers or crates for distribution and, in some embodiments, in which the produce is sold to consumers). The location of the tags 402 may be tracked as described above. Then, during a food-borne illness outbreak, the stricken person information may be cross-referenced against the location tracking information in an attempt to identify a correspondence between the sickened people and a food product from a group or category of suspected food products, a food product that had an end distribution pattern near the locations of the sickened persons, or a food product classified in some other manner. In this manner, identification of the culprit food product may be identified rapidly. It is contemplated that culprit product identification may be made faster than if conventional analysis is made.

Once the culprit food product is identified, the food product may be recalled. The tracking information may be used both downstream and upstream to facilitate product recall and other remedial actions. For example, the site at which the pathogen was introduced may be identified and the pathogen may be eradicated. Also, the last detected location of food units that might be contaminated and/or subject to recall may be identified. If those items are still at grocers or restaurants, the grocer or restaurant may be alerted and the food may be pulled from sale or use. Also, for product that was purchased by a consumer, the specific purchaser of some of the items may be identified and contacted using records establishing a correlation between purchaser and tagged object. In some embodiments, return signals may be analyzed to identify the present location of recalled units and action may be taken to retrieve those units from restaurants, homes, grocers or other locations.

Another example application is the tracking of items that are due for service, or product upgrade or recall. An exemplary embodiment of a product recall with respect to a car will be described, but modifications to the method for situations involving routine service of product upgrade will be apparent without further explanation. In this embodiment, the probe P emits a guided surface wave that illuminates tags P associated with cars. Receivers 408 are positioned along roadways, parking areas, driveways or other locations that cars may pass. As a car passes one of the receivers 408, the return signal from the associated tag 402 will be received by the receiver 408. The tag identifier or vehicle data associated with the tag identifier, such as a vehicle identification number (VIN), may be cross-referenced against a database that stores which cars, by make and model, have completed necessary work to address a product safety recall. Data regarding completion of recall work may be obtained from car dealers and other service providers as the work is performed. If the vehicle is determined to have completed the work, no additional action may be taken. If the vehicle is determined to have not completed the work, additional action may be taken. For instance, data may be transmitted to the tag 402 via an encoded carrier message in the guided surface wave. The data may prompt the tag 402 to interface with electronics of the vehicle to display a message to the driver that there is a product recall that should be addressed. Other actions may include attempting to contact an owner of the vehicle or an enforcement authority by phone, email, text or data message, convention mail, etc.

Another application may be charging a driver or vehicle owner for use of a toll road. In this example, receivers 408 may be positioned at the entrances and exits from the toll road, or along the toll road. As return signals from tags 402 associated with the vehicles or drivers that pass the receivers 408 are received, appropriate charges may be made against an account or credit card that has been previously associated with the driver or vehicle in the computer system 418.

In another embodiment, return signals from tags 402 or the lack of a return signal may be used to identify counterfeit goods or authenticate legitimate goods. In one exemplary approach, each legitimate object is associated with a tag 402 having a unique identifier. At various times, the tag identifier may be checked against a database of tag identifiers that are known to be associated with legitimate goods. Exemplary times at which goods are checked may include at the time of passing through a customs control checkpoint and when possession or title in the goods are transferred between parties (e.g., from manufacturer to importer, from importer to distributor, from distributor to store owner, from store owner to consumer). If there is a match between the received tag identifier and the database of known legitimate tag identifiers, the goods may be cleared by the customs authority or accepted by the receiving party. If there is not a match or no tag 402 is present, the customs authority may confiscate the goods and perform an investigation or the receiving party may reject the goods.

As is evident from the foregoing example, the amount of tracking and data collected with respect to various objects 404 will depend on the degree of interest in the objects 404 and the reason for tracking the object 404. Beyond tracking of an object 404, the associated tag 402 may be used for additional purposes. Examples will be provided. In these examples, data may be transferred to the tag 402 or queries or commands may be transmitted to the tag 402. In these situations, the data, query or command may be transmitted by way of a communication link between the tag 402 and a receiver 404 or may be encoded in a message addressed to the tag 402 and forming part of the guided surface wave (e.g., as an encoded carrier message).

In one embodiment, data in addition to the tag identifier may be stored by the tag 402. The stored data, or selected elements of the stored data, may be transmitted as part of an automated return signal. In other situations, the stored data, or selected elements of the stored data, may be transmitted in a signal responsive to a query or command. Information stored by the tag 402 may change over time as is appropriate to support operational functionality. Stored data elements may include, but are not limited to, locations at which presence of the tag 402 was previously determined (e.g., a location history record); identifiers of receivers 408 that received a return signal from the tag 402; an identity or location of the manufacturer, importer, distributor or owner of the object 404 that is associated with the tag 402; time and date of manufacture, packaging or other processing; an association of one or more additional objects 404 with the tag 402; customs clearance data; location, time and date, and/or other details related to certain events such as crossing a port of entry, manufacture, purchase, purchase amount, etc.; a product expiration date; a version number or value; product functions; a website or other data store from which more product information, warranty information, legal terms, or intellectual property coverage information is available; information about obtaining product support or ordering accessories or replacement parts; etc.

In various embodiments, the guided surface wave will be present over long periods of time to illuminate tags 402 over large geographic areas. At some point, the value of a return signal for certain products may no longer be of interest to one or more parties. For example, a purchaser of an object 404 may not wish to have the object's tag 402 send return signals due to privacy concerns. As another example, after food is consumed, a tag 402 associated with the food's packaging is of little value. In these situations, it may be possible to recycle tags, destroy tags, turn the return signal feature of the tags off, contact a tracking data system (e.g., the computer system 418) and opt out of further tracking of tags, or other action that alters the operation of the tags, receivers or computer system.

In one embodiment, tags 402 may be responsive to guided surface waves of more than one frequency. For instance, a tag 402 may emit a first return signal when in the presence of a guided surface wave of a first frequency that is produced by a first probe and covers a widespread area as described in connection with FIG. 25 and may emit a second return signal when in the presence of a guided surface wave of a second frequency that is produced by a second probe and covers a local area (e.g., an area corresponding to specific site) as described in connection with FIG. 24. The return signal responsive to the wide-area guided surface wave of the first frequency may be at a frequency that is different than the frequency of the return signal of the local-area guided surface wave of the second frequency. In this manner, the return signals may be distinguished and/or received by different receivers 408.

2(G). Computer System

The computer system in the various embodiments may be any appropriate system, such a personal computer, a server or a distributed system (e.g., a “cloud” computing environment). With additional reference to FIG. 26, an exemplary computer system 418 communicatively coupled with a receiver 408 is illustrated. If appropriate, the computer system 418 may communicate with plural receivers 408. If applicable, the computer system 418 may have operable communication with one or more probes P to control when the probes 300 generate a guided surface wave and characteristics of the guided surface waves, and to control the probes 300 to include data or commands for transmission to one or more tags 402 in the guided surface waves.

The computer system 418, together with the receivers 408, probes P and tags 402, may carry out the techniques that are described in this disclosure. As indicated, the computer system 418 communicates with the receiver 408 over any appropriate communications medium 420. In addition to carrying out the operations described herein, the computer system 418 may be a central registration system or some other form of management platform to manage the logical association of tags 402 with objects 404.

The computer system 418 may be implemented as a computer-based system that is capable of executing computer applications (e.g., software programs), including a tag management function 448 that, when executed, carries out functions of the computer system 418 that are described herein. The tag management function 448 and a database 450 may be stored on a non-transitory computer readable medium, such as a memory 452. The database 450 may be used to store various information sets used to carry out the functions described in this disclosure. The memory 452 may be a magnetic, optical or electronic storage device (e.g., hard disk, optical disk, flash memory, etc.), and may comprise several devices, including volatile and non-volatile memory components. Accordingly, the memory 452 may include, for example, random access memory (RAM) for acting as system memory, read-only memory (ROM), solid-state drives, hard disks, optical disks (e.g., CDs and DVDs), tapes, flash devices and/or other memory components, plus associated drives, players and/or readers for the memory devices.

To execute logical operations, the computer system 418 may include one or more processors 454 used to execute instructions that carry out logic routines. The processor 454 and the memory 452 may be coupled using a local interface 456. The local interface 456 may be, for example, a data bus with accompanying control bus, a network, or other subsystem.

The computer system 418 may have various input/output (I/O) interfaces for operatively connecting to various peripheral devices. The computer system 418 also may have one or more communications interfaces 458. The communications interface 458 may include for example, a modem and/or a network interface card. The communications interface 458 may enable the computer system 418 to send and receive data signals to and from other computing devices, the receivers 408 and the probes P via the communications medium 420. In particular, the communications interface 458 may operatively connect the computer system 418 to the communications medium 420.

The receiver 408 includes communications circuitry, such as radio circuitry 460 to receive return signals from the tags 402 and a communications interface 462 to establish operable communications with other devices over the communications medium 420. The radio circuitry 460 may include one or more antennas and radio receivers (or transceivers in the case where the receiver 408 transmits data or commands to the tags 402).

Overall functionality of the receiver 408 may be controlled by a control circuit 464 that includes, for example a processing device for executing logical instructions. The receiver 408 also may include a memory 466 for storing data and the logical instructions in the form of executable code. The memory 466 may be a non-transitory computer readable medium such as one or more of a buffer, a flash memory, a hard drive, a removable media, a volatile memory, a non-volatile memory, a random access memory (RAM), or other suitable device. In a typical arrangement, the memory 466 includes a non-volatile memory for long term data storage and a volatile memory that functions as system memory for the control circuit 464. The receiver 408 may include any other appropriate components such as, but not limited to, a display, a speaker, a microphone, a user interface (e.g., a keypad and/or a touch-sensitive input), motion sensors, location determining elements (e.g., a GPS receiver), etc.

3. Conclusion

Features that are described and/or illustrated with respect to one embodiment may be used in the same way or in a similar way in one or more other embodiments and/or in combination with or instead of the features of the other embodiments. Therefore, any one disclosed feature may be combinable or interchangeable with any other features.

Furthermore, although certain embodiments have been shown and described, it is understood that equivalents and modifications falling within the scope of the appended claims will occur to others who are skilled in the art upon the reading and understanding of this specification. 

1. An object identification system, comprising: a guided surface waveguide probe comprising a charge terminal elevated over a terrestrial medium that produces a guided surface wave by generating at least one resultant field that synthesizes a wave front incident at a complex Brewster angle of incidence (θ_(i,B)) of the terrestrial medium; and an object identification tag comprising a receive structure and a tag circuit, the tag circuit coupled to the receive structure and electrically powered as a load on the probe by conversion of the guided surface wave to electrical current at the receive structure, the tag circuit configured to emit a return signal containing a tag identifier when electrically powered by presence of the guided surface wave.
 2. The system of claim 23, wherein the guided surface waveguide probe comprises a charge terminal elevated over a terrestrial medium configured to generate at least one resultant field that synthesizes a wave front incident at a complex Brewster angle of incidence (θ_(i,B)) of the terrestrial medium.
 3. The system of claim 1, wherein the system further comprises a receiver that receives the return signal.
 4. The system of claim 3, further comprising a computer system operatively coupled to the receiver and that determines the tag identifier and identifies an object that was previously associated with the tag.
 5. The system of claim 3, wherein the tag is addressable and the receiver transmits an addressed message to the tag.
 6. The system of claim 1, wherein the guided surface wave has a first frequency and the return signal has a second frequency higher than the first frequency.
 7. The system of claim 1, wherein the return signal is a radio frequency (RF) signal.
 8. (canceled)
 9. The system of claim 23, wherein one or more tags are queried to each return data in addition to the tag identifier.
 10. The system of claim 23, wherein one or more tags are commanded to not transmit respective return signals for at least a period of time.
 11. The system of claim 23, wherein a size of the illumination area is controlled to be non-overlapping with an adjacent second illumination area that is defined by a second guided surface wave produced by a second guided surface waveguide probe.
 12. The system of claim 23, wherein a shape of the illumination area is controlled by producing the guided surface wave to vary as a function of direction.
 13. The system of claim 23, wherein the illumination area is global.
 14. The system of claim 1, wherein the guided surface wave penetrates high permittivity material to power tags located therein.
 15. The system of claim 1, wherein the tag is addressable and the guided surface wave includes an addressed message for the tag.
 16. The system of claim 1, wherein the tag is addressable and, upon receipt of a message addressed to the tag, the tag circuit executes logic to carry out a function corresponding to the message.
 17. The system of claim 16, wherein the function includes reading data from a memory of the tag and transmitting a signal containing the read data.
 18. The system of claim 16, wherein the function includes storing data in a memory of the tag.
 19. The system of claim 16, wherein the function is to not transmit a return signal for at least a period of time.
 20. The system of claim 16, wherein the message is sent by a receiver configured to receive the return signal.
 21. The system of claim 1, wherein the system further comprises a computer system, the computer system configured to manage an inventory of objects for a site, each object associated with a tag and the inventory managed according to tag identifiers in return signals that are respectively received from the tags.
 22. The system of claim 1, wherein the system further comprises a computer system, the computer system configured to identify locations of objects, each object associated with a tag and the locations determined according to return signals respectively received from the tags.
 23. An object identification system, comprising: a guided surface waveguide probe that produces a guided surface wave having a frequency-dependent illumination area in which tags responsive to a frequency of the guided surface wave are powered and emit respective return signals; and an object identification tag comprising a receive structure and a tag circuit, the tag circuit coupled to the receive structure and electrically powered as a load on the probe by conversion of the guided surface wave to electrical current at the receive structure, the tag circuit configured to emit a return signal containing a tag identifier when electrically powered by presence of the guided surface wave. 